Wireless communications system and wireless digital receiver for use therein

ABSTRACT

A base station  2  demodulates a wireless signal after downconverting the signal to a low-frequency signal whose center frequency is fi [Hz] and oversampling the signal. A mobile station  3  demodulates a wireless signal after downconverting the signal to a low-frequency signal whose center frequency is fd [Hz] and undersampling the signal. The same sampling frequency fs [Hz] is used in the base station  2  and in the mobile station  3 . The sampling frequency fs [Hz] is set to a value that is an even-number multiple of a wireless symbol transmission rate such that oversampling is done in the base station  2  and undersampling is done in the mobile station  3.  The center frequency fi [Hz] is ½ to 1 times a frequency corresponding to the bandwidth and is ½ N  (N is a natural number) times the sampling frequency fs [Hz].

TECHNICAL FIELD

The present invention relates to a wireless communications system and awireless digital receiver for use therein, and more particularly to awireless communications system using an FDD (Frequency Division Duplex)architecture and a wireless digital receiver for use therein.

BACKGROUND ART

One of the conventional wireless communications systems using the FDDarchitecture is a wireless communications system using a DSRC (DedicatedShort Range Communications) architecture (hereinafter referred to as a“DSRC system”).

The DSRC system standard specifies that when transmitting a firstwireless signal from a first wireless communications device provided onthe road (hereinafter referred to as a “base station”) to a secondwireless communications device provided in a vehicle (hereinafterreferred to as a “mobile station”) (hereinafter such a transmission willbe referred to as a “downlink”), one of 5775 [MHz], 5780 [MHz], 5785[MHz], 5790 [MHz], 5795 [MHz], 5800 [MHz] and 5805 [MHz] is used as thecenter frequency.

The DSRC system standard also specifies that when transmitting a secondwireless signal from the mobile station to the base station (hereinaftersuch a transmission will be referred to as an “uplink”), a centerfrequency that is away from that used for the downlink by 40.000 [MHz]is used. Specifically, if 5775 [MHz] is used as the center frequency forthe downlink, 5815 [MHz] is used as the center frequency for the uplink.Similarly, if 5780 [MHz] is used for the downlink, 5820 [MHz] is usedfor the uplink. If 5785 [MHz] is used for the downlink, 5825 [MHz] isused for the uplink. If 5790 [MHz] is used for the downlink, 5830 [MHz]is used for the up link. If 5795 [MHz] is used for the downlink, 5835[MHz] is used for the uplink. If 5800 [MHz] is used for the downlink,5840 [MHz] is used for the uplink. If 5805 [MHz] is used for thedownlink, 5845 [MHz] is used for the uplink.

In a section on the technical requirements for wireless equipment inDSRC system standard specifications, standards for image response arespecified only for the base station.

Where a demodulation process is performed by a digital signal processingcircuit, in order to convert a received modulated high-frequency signalto a frequency suitable for the digital signal processing circuit, afrequency conversion circuit for downconverting the modulatedhigh-frequency signal needs to be provided preceding the digital signalprocessing circuit.

In view of the technical requirements for wireless equipment, it ispreferred that a frequency conversion circuit employing a LOW-IFarchitecture, for example, is used in the base station. This is becauseit is possible with the LOW-IF architecture to remove an imagedisturbing signal without using an image suppression filter in ahigh-frequency part, as described in Non-Patent Document 1 (J. Crols andMichiel S. J. Steyaert, “Low-IF Topologies for High-Performance AnalogFront Ends of Fully Integrated Receivers”, IEEE TRANSACTIONS ON CIRCUITSAND SYSTEMS II: ANALOG AND DIGITAL SIGNAL PROCESSING, VOL. 45, NO. 3,March 1998).

As described in Non-Patent Document 1, with the LOW-IF architecture, thecenter frequency of a received modulated high-frequency signal isdownconverted to a frequency that is about a few times as great as thesignal bandwidth of the modulated high-frequency signal. Then, thedownconverted signal is directly sampled by a sampler and demodulated bya digital signal processing circuit. The LOW-IF architecture isadvantageous in that it offers better reception characteristics and highdegrees of integration.

For the mobile station, however, no image response standard isspecified. Therefore, it is possible to use a frequency converter inwhich a local oscillator is shared by the transmitter and the receiver.Thus, with the mobile station, a single-conversion architecture can beemployed. Therefore, the mobile station can be provided at a low cost.

As described above, where a frequency converter employing the LOW-IFarchitecture is used in the base station, a frequency-converted signalis converted into a signal having a frequency that is about a few timesas great as the signal bandwidth of the received modulatedhigh-frequency signal.

Where the mobile station uses a frequency converter employing thesingle-conversion architecture in which a local oscillator is shared bythe transmitter and the receiver, a frequency-converted signal isconverted into a signal having a frequency that is equal to thedifference between the uplink and the downlink. Typically, thefrequencies are different from each other. This is illustrated in FIG.20 to FIG. 22.

FIG. 20 is a diagram schematically showing a conventional base station9000 and a conventional mobile station 9001 communicating with eachusing the DSRC system. In FIG. 20, a frequency fc denotes the centerfrequency for the uplink, and the value thereof is one of 5815 [MHz],5820 [MHz], 5825 [MHz], 5830 [MHz], 5835 [MHz], 5840 [MHz] and 5845[MHz]. Moreover, in FIG. 20, a frequency fd denotes the differencebetween the center frequency of the signal used for the uplink and thatof the signal used for the downlink, and the value thereof is 40.000[MHz]. As shown in FIG. 20, a signal is uplinked from the mobile station9001 to the base station 9000 with the center frequency fc. A signal isdownlinked from the base station 9000 to the mobile station 9001 with acenter frequency fc-fd. In the DSRC system, it is specified that thechannel bandwidth is 5 [MHz].

FIG. 21 is a diagram showing a general configuration of a conventionalbase-station wireless communications device employing the LOW-IFarchitecture. FIG. 22 is a diagram showing a general configuration of aconventional mobile-station wireless communications device employing thesingle-conversion architecture. For the purpose of simplifying theproblem, the following description will only discuss thesignal-receiving operation at the mobile-station wireless communicationsdevice and the base-station wireless communications device.

First, referring to FIG. 20 and FIG. 21, the signal-receiving operationat the base-station wireless communications device will be described. InFIG. 21, the base-station wireless communications device includes anantenna 9200, a band-pass filter 9216, a transmission/reception selectorswitch 9211, an amplifier 9201, a first mixer 9202, a second mixer 9203,a first local oscillator 9206, a first low-pass filter 9204, a secondlow-pass filter 9205, a first sampler 9207, a second sampler 9208, asampling signal generator 9209, a demodulation digital circuit 9210, atransmission high-frequency circuit 9212, a third mixer 9213, a secondlocal oscillator 9214 and a transmitter circuit 9215.

In the base-station wireless communications device, the signal-receivingoperation is performed by using the antenna 9200, the band-pass filter9216, the transmission/reception selector switch 9211, the amplifier9201, the first mixer 9202, the second mixer 9203, the first localoscillator 9206, the first low-pass filter 9204, the second low-passfilter 9205, the first sampler 9207, the second sampler 9208, thesampling signal generator 9209 and the demodulation digital circuit9210.

In the signal-receiving operation, the transmission/reception selectorswitch 9211 is switched so that the antenna 9200 and the amplifier 9201are connected to each other. A modulated high-frequency signal R(t) fromthe mobile station 9001 received by the antenna 9200 whose centerfrequency is fc is inputted to the amplifier 9201. The amplifier 9201amplifies the modulated high-frequency signal R(t) to an appropriatelevel, and inputs the amplified signal to the first mixer 9202 and thesecond mixer 9203. The first local oscillator 9206 outputs a sine wavewhose center frequency is fc-fa. As described in Non-Patent Document 1,it is preferred that fa is a frequency that is about a few times asgreat as the channel bandwidth of the modulated high-frequency signalR(t).

The first mixer 9202 multiplies the sine wave outputted from the firstlocal oscillator 9206 whose center frequency is fc-fa with the modulatedhigh frequency signal R(t) to output a modulatedlow-to-intermediate-frequency signal in-phase component RXI(t) whosecenter frequency is fa.

The second mixer 9203 multiplies a signal outputted from the first localoscillator 9206 whose center frequency is fc-fa and whose phase isshifted from that of the sine wave by π/2 with the modulatedhigh-frequency signal R(t) to output a modulatedlow-to-intermediate-frequency signal quadrature component RXQ(t) whosecenter frequency is fa.

The first sampler 9207 samples the modulatedlow-to-intermediate-frequency signal in-phase component RXI(t) insynchronism with a signal outputted from the sampling signal generator9209 whose frequency is fs1 to output an in-phase component sampledsignal I(mTs1).

The second sampler 9208 samples the modulatedlow-to-intermediate-frequency signal quadrature component RXQ(t) insynchronism with a signal outputted from the sampling signal generator9209 whose frequency is fs1 to output a quadrature component sampledsignal Q(mTs1).

Herein, m is an integer, and Ts1 is the inverse of the sampling signalfrequency fs1, .e., Ts1=1/fs1. In order to facilitate the signalprocessing operation at the demodulation digital circuit 9210, fs1 is inmany cases set to a value that is equal to fa multiplied by 2^(N) (N isa natural number: N=1, 2, 3, . . . ).

The demodulation digital circuit 9210 receives the in-phase componentsampled signal I(mTs1) and the quadrature component sampled signalQ(mTs1) as input signals, and demodulates the signals to output receiveddata after removing the image disturbing signal, as described inNon-Patent Document 1.

Next, referring to FIG. 21 and FIG. 22, the signal-receiving operationat the mobile-station wireless communications device will be described.In FIG. 22, the mobile-station wireless communications device includesan antenna 9100, a band-pass filter 9112, a transmission/receptionselector switch 9108, an amplifier 9101, a first mixer 9102, a localoscillator 9103, a low-pass filter 9104, a sampler 9105, a samplingsignal generator 9106, a demodulation digital circuit 9107, atransmission high-frequency circuit 9109, a second mixer 9110 and atransmitter circuit 9111.

In the mobile-station wireless communications device, thesignal-receiving operation is performed by using the antenna 9100, theband-pass filter 9112, the transmission/reception selector switch 9108,the amplifier 9101, the first mixer 9102, the local oscillator 9103, thelow-pass filter 9104, the sampler 9105, the sampling signal generator9106 and the demodulation digital circuit 9107.

In the signal-receiving operation, the transmission/reception selectorswitch 9108 is switched so that the antenna 9100 and the amplifier 9101are connected to each other. A modulated high-frequency signal RL(t)from the base station 9000 received by the antenna 9100 whose centerfrequency is fc-fd is first passed through the band pass filter 9112 toremove signals of frequency bands that are used neither in the basestation nor in the mobile station, and is then inputted to the amplifier9101. The amplifier 9101 amplifies the modulated high-frequency signalRL(t) to an appropriate level, and inputs the amplified signal to thefirst mixer 9102. The first local oscillator 9103 outputs a sine wavewhose center frequency is fc.

The first mixer 9102 multiplies the sine wave outputted from the localoscillator 9103 whose center frequency is fc with the modulatedhigh-frequency signal RL(t) to output a modulatedlow-to-intermediate-frequency signal L(t) whose center frequency is fdto the low-pass filter 9104.

In the frequency conversion at the first mixer 9102, a signal whosecenter frequency is fc+fd is an image disturbing signal. However, sincethe image response is not specified in the technical requirements forwireless equipment used in the mobile station in the DSRC systemstandard, a lower-order, inexpensive low-pass filter can be used as thefilter following the first mixer 9102. If the image disturbing signalwere to be a problem, signal components of only the necessary bands canbe extracted by using a complex filter.

The sampler 9105 samples the modulated low-to-intermediate-frequencysignal L(t) outputted from the low-pass filter 9104 whose centerfrequency is fd in synchronism with a signal outputted from the samplingsignal generator 9106 whose frequency is fs2 output a sampled signalLs(mTs2). Herein, m is an integer, and Ts2 is a value represented by theinverse (1/fs2) of the sampling signal frequency fs2. In order tofacilitate the signal processing operation at the demodulation digitalcircuit 9107, fs2 is in many cases set to a value that is equal to fdmultiplied by 2^(N) (N is a natural number: N=1, 2, 3, . . . )

The demodulation digital circuit 9107 receives the sampled signalLs(mTs2) as an input signal, and demodulates the signal to outputreceived data.

Other background art publications related to the present inventioninclude Non-Patent Document 2 (Mikko Valkama, et al., “Advanced Methodsfor I/Q Imbalance Compensation in Communication Receivers” IEEETRANSACTIONS ON SIGNAL PROCESSING, Vol. 49, No. 10, pp. 2335-2344,October 2001) and Non-Patent Document 3 (Kiyomichi Araki ed., “SoftwareMusen No Kiso To Oyo (Basics and Applications of Software Radio)”, SIPECCorporation Knowledge Service Department, p. 123, October 2002).

As described above, in the base station employing the LOW-IFarchitecture, the center frequency fa of the signals RXI(t) and RXQ(t)inputted to the first and second samplers 9207 and 9208 is about a fewtimes as great as the signal bandwidth of the modulated high-frequencysignal R(t). In the mobile station employing the single-conversionarchitecture, the center frequency fd of the signal L(t) inputted to thesampler 9105 is equal to the difference (40.000 [MHz]) between theuplink frequency and the downlink frequency as specified in the DSRCsystem standard.

Therefore, the center frequency of the signal inputted to the sampler inthe mobile station is substantially different from that of the signalinputted to the sampler in the base station, whereby the frequency ofthe sampling signal used in the sampler 9105 in the mobile station isdifferent from that of the sampling signal used in the first and secondsamplers 9207 and 9208 in the base station.

Therefore, with the conventional system, the sampling frequency for thebase station and that for the mobile station need to be set to differentvalues even though their demodulation digital circuits are substantiallythe same in function. Thus, it is necessary to provide two differentdemodulation digital circuits for the base station and for the mobilestation. Although it is desirable to realize a common demodulationdigital circuit for the base station and for the mobile station in orderto provide an inexpensive transceiver, it is difficult to realize such acommon demodulation digital circuit for reasons stated above.

DISCLOSURE OF THE INVENTION

Therefore, an object of the present invention is to realize a wirelesscommunications system in which a common sampling frequency is used bythe base station and the mobile station, thereby providing wirelessdigital receivers for the base station and for the mobile station at alow cost and reducing the overall cost of the wireless communicationssystem.

The present invention has the following features to attain the objectmentioned above.

A first aspect of the present invention is directed to a wirelesscommunications system for transmitting/receiving a first wireless signalfrom a first wireless communications device and a second wireless signalfrom a second wireless communications device, the first and secondwireless signals having different frequency bands from each other,wherein: the first wireless communications device includes: a firstfrequency converter operable to downconvert the second wireless signaltransmitted from the second wireless communications device to a firstlow-frequency signal; a first sampler operable to oversample the firstlow-frequency signal downconverted by the first frequency converter; anda first demodulation digital circuit operable to demodulate the signaloversampled by the first sampler; the signal demodulated by the firstdemodulation digital circuit has a center frequency of fi [Hz]; thesecond wireless communications device includes: a second frequencyconverter operable to downconvert the first wireless signal transmittedfrom the first wireless communications device to a second low-frequencysignal whose center frequency fd [Hz] is equal to a difference between acenter frequency of the first wireless signal and that of the secondwireless signal; a second sampler operable to undersample the secondlow-frequency signal downconverted by the second frequency converter;and a second demodulation digital circuit operable to demodulate thesignal undersampled by the second sampler; a sampling frequency used inthe first sampler and that used in the second sampler are the samesampling frequency fs [Hz]; the sampling frequency fs [Hz] is set to avalue that is an even-number multiple of a wireless symbol transmissionrate such that oversampling is done in the first sampler andundersampling is done in the second sampler; and the center frequency fi[Hz] is ½ to 1 times a frequency corresponding to a bandwidth of thefirst and second wireless signals and is ½^(N) (N is a natural number)times the sampling frequency fs [Hz].

In a preferred embodiment, where the bandwidth of the first and secondwireless signals is 2×Bch [Hz] and the wireless symbol transmission rateis fsym [Hz], the sampling frequency fs [Hz] and the center frequency fi[Hz] are expressed as shown in the following expressions:$\begin{matrix}{{fi} = \frac{2{kfsym}}{2^{N}}} \\{{fs} = {2^{N}{fi}}}\end{matrix}$where k is an integer satisfying $\begin{matrix}{{\frac{{fd} + {Bch}}{\left( {n + 1} \right){fsym}} \leqq k \leqq \frac{{fd} - {Bch}}{n\quad{fsym}}}{and}} & {{Exp}.\quad 12} \\{k \leqq \frac{fd}{2{fsym}}} & {{Exp}.\quad 14}\end{matrix}$and N is an integer satisfying $\begin{matrix}{{\log\quad 2\left\{ \frac{{fd} + {bch}}{\left( {n + 1} \right){Bch}} \right\}} \leqq N \leqq {\log\quad 2\left\{ \frac{2\left( {{fd} - {Bch}} \right)}{n\quad{Bch}} \right\}}} & {{Exp}.\quad 22}\end{matrix}$where n is an integer satisfying $\begin{matrix}{1 \leqq n \leqq {\frac{{fd} - {Bch}}{2{Bch}}.}} & {{Exp}.\quad 7}\end{matrix}$

In a preferred embodiment, the first frequency converter downconvertsthe second wireless signal transmitted from the second wirelesscommunications device to a first low-frequency signal whose centerfrequency is fj [Hz]; and the first low-frequency signal is demodulatedby the first demodulation digital circuit after being corrected to asignal whose center frequency is fi [Hz] at a position preceding orfollowing the first sampler.

In a preferred embodiment, the center frequency fd is 40.000 [MHz]; andthe frequency fi and the sampling frequency fs are fi=3.072 [MHz] andfs=24.576 [MHz], fi=3.072 [MHz] and fs=12.288 [MHz], fi=4.608 [MHz] andfs=36.864 [MHz], fi=4.096 [MHz] and fs=32.768 [MHz], or fi=3.584 [MHz]and fs=28.672 [MHz].

In a preferred embodiment, the first demodulation digital circuitincludes: a first quadrature demodulator operable toquadrature-demodulate the signal over sampled by the first sampler; afirst low-pass filter operable to low-pass-filter the signal quadraturedemodulated by the first quadrature demodulator; and a first receiveddata reproducing section operable to reproduce received data from thesignal low-pass-filtered by the first low-pass filter; the seconddemodulation digital circuit includes: a second quadrature demodulatoroperable to quadrature-demodulate the signal undersampled by the secondsampler; a second low-pass filter operable to low-pass-filter the signalquadrature-demodulated by the second quadrature demodulator; and asecond received data reproducing section operable to reproduce receiveddata from the signal low-pass-filtered by the second low-pass filter;the first quadrature demodulator converts the signal oversampled by thefirst sampler to a signal including a component whose center frequencyis zero; and the second quadrature demodulator converts the signalundersampled by the second sampler to a signal including a componentwhose center frequency is zero.

In a preferred embodiment, the first demodulation digital circuitincludes: a first complex filter operable to filter, by using a digitalfilter, either one of a positive frequency component and a negativefrequency component of the signal oversampled by the first sampler whosecenter frequency is closer to zero; and a first received datareproducing section operable to reproduce received data from the signalfiltered by the first complex filter; and the second demodulationdigital circuit includes: a second complex filter operable to filter, byusing a digital filter, either one of a positive frequency component anda negative frequency component of the signal under sampled by the secondsampler whose center frequency is closer to zero; and a second receiveddata reproducing section operable to reproduce received data from thesignal filtered by the second complex filter.

In a preferred embodiment, the first demodulation digital circuitincludes: a first quadrature demodulator operable toquadrature-demodulate the signal over sampled by the first sampler; afirst low-pass filter operable to low-pass-filter the signal outputtedfrom the first quadrature demodulator; and a first received datareproducing section operable to reproduce received data from the signallow-pass-filtered by the first low-pass filter; the second demodulationdigital circuit includes: a second quadrature demodulator operable toquadrature-demodulate the signal under sampled by the second sampler; asecond low-pass filter operable to low-pass-filter the signalquadrature-demodulated by the second quadrature demodulator; and asecond received data reproducing section operable to reproduce receiveddata from the signal low-pass-filtered by the second low-pass filter;the first quadrature demodulator converts the signal oversampled by thefirst sampler to a signal including a component whose center frequencyis zero; and the second quadrature demodulator converts the signalundersampled by the second sampler to a signal including a componentwhose center frequency is zero.

In a preferred embodiment, the frequency fj [Hz] is 3.000 [MHz].

A second aspect of the present invention is directed to a wirelessdigital receiver in a wireless communications system fortransmitting/receiving a first wireless signal from a first wirelesscommunications device and a second wireless signal from a secondwireless communications device, the first and second wireless signalshaving different frequency bands from eachother, the wireless digitalreceiver receiving the second wireless signal in the first wirelesscommunications device and digitally demodulating the second wirelesssignal, the wireless digital receiver including: a frequency converteroperable to downconvert the second wireless signal transmitted from thesecond wireless communications device to a low-frequency signal whosecenter frequency is fi [Hz]; a sampler operable to oversample thelow-frequency signal downconverted by the frequency converter; and ademodulation digital circuit operable to demodulate the signaloversampled by the sampler, wherein: a sampling frequency used in thesampler and that used in the second wireless communications device arethe same sampling frequency fs [Hz]; the sampling frequency fs [Hz] isset to a value that is an even-number multiple of a wireless symboltransmission rate such that oversampling is done in the sampler andundersampling is done in a sampler of the second wireless communicationsdevice; and the center frequency fi [Hz] of the low-frequency signal is½ to 1 times a frequency corresponding to a bandwidth of the first andsecond wireless signals and is ½^(N) (N is a natural number) times thesampling frequency fs [Hz].

In a preferred embodiment, where the bandwidth of the first and secondwireless signals is 2×Bch [Hz] and the wireless symbol transmission rateis fsym [Hz], the sampling frequency fs [Hz] and the center frequency fi[Hz] of the low-frequency signal are expressed as shown in the followingexpressions: $\begin{matrix}{{fi} = \frac{2{kfsym}}{2^{N}}} \\{{fs} = {2^{N}{fi}}}\end{matrix}$where k is an integer satisfying $\begin{matrix}{\frac{{fd} + {Bch}}{\left( {n + 1} \right){fsym}} \leqq k \leqq \frac{{fd} - {Bch}}{n\quad{fsym}}} & {{Exp}.\quad 12} \\{and} & \quad \\{k \leqq \frac{fd}{2{fsym}}} & {{Exp}.\quad 14}\end{matrix}$and N is an integer satisfying $\begin{matrix}{{\log\quad 2\left\{ \frac{{fd} + {bch}}{\left( {n + 1} \right){Bch}} \right\}} \leqq N \leqq {\log\quad 2\left\{ \frac{2\left( {{fd} - {Bch}} \right)}{n\quad{Bch}} \right\}}} & {{Exp}.\quad 22}\end{matrix}$where n is an integer satisfying $\begin{matrix}{1 \leqq n \leqq {\frac{{fd} - {Bch}}{2{Bch}}.}} & {{Exp}.\quad 7}\end{matrix}$

In a preferred embodiment, the center frequency fi and the samplingfrequency fs are fi=3.072 [MHz] and fs=24.576 [MHz], fi=3.072 [MHz] andfs=12.288 MHz], fi=4.608 [MHz] and fs=36.864 [MHz], fi=4.096 [MHz] andfs=32.768 [MHz], or fi=3.584 [MHz] and fs=28.672 [MHz].

In a preferred embodiment, the demodulation digital circuit includes: aquadrature demodulator operable to quadrature-demodulate the signaloversampled by the sampler; a low-pass filter operable tolow-pass-filter the signal quadrature-demodulated by the quadraturedemodulator; and a received data reproducing section operable toreproduce received data from the signal low-pass-filtered by thelow-pass filter; and the quadrature demodulator converts the signaloversampled by the sampler to a signal including a component whosecenter frequency is zero.

In a preferred embodiment, the demodulation digital circuit includes: acomplex filter operable to filter, by using a digital filter, either oneof a positive frequency component and a negative frequency component ofthe signal oversampled by the sampler whose center frequency is closerto zero; and a received data reproducing section operable to reproducereceived data from the signal filtered by the complex filter.

A third aspect of the present invention is directed to a wirelessdigital receiver in a wireless communications system fortransmitting/receiving a first wireless signal from a first wirelesscommunications device and a second wireless signal from a secondwireless communications device, the first and second wireless signalshaving different frequency bands from each other, the wireless digitalreceiver receiving the first wireless signal in the second wirelesscommunications device and digitally demodulating the first wirelesssignal, the wireless digital receiver including: a frequency converteroperable to downconvert the first wireless signal transmitted from thefirst wireless communications device to a low-frequency signal whosecenter frequency fd [Hz] is equal to a difference between a centerfrequency of the first wireless signal and that of the second wirelesssignal; a sampler operable to undersample the low-frequency signaldownconverted by the frequency converter; and a demodulation digitalcircuit operable to demodulate the signal undersampled by the sampler,wherein: a sampling frequency used in the sampler and that used in thefirst wireless communications device are the same sampling frequency fs[Hz]; and the sampling frequency fs [Hz] is set to a value that is aneven-number multiple of a wireless symbol transmission rate such thatundersampling is done in the sampler and oversampling is done in asampler of the first wireless communications device.

In a preferred embodiment, where the bandwidth of the first and secondwireless signals is 2×Bch [Hz] and the wireless symbol transmission rateis fsym [Hz], the sampling frequency fs [Hz] is expressed as shown inthe following expression:fs=2kfsymwhere k is an integer satisfying $\begin{matrix}{{\frac{{fd} + {Bch}}{\left( {n + 1} \right){fsym}} \leqq k \leqq \frac{{fd} - {Bch}}{n\quad{fsym}}}{and}} & {{Exp}.\quad 12} \\{k \leqq \frac{fd}{2{fsym}}} & {{Exp}.\quad 14}\end{matrix}$where n is an integer satisfying $\begin{matrix}{1 \leqq n \leqq {\frac{{fd} - {Bch}}{2{Bch}}.}} & {{Exp}.\quad 7}\end{matrix}$

In a preferred embodiment, the center frequency fd is 40.000 [MHz]; andthe sampling frequency fs is 24.576 [MHz], 12.288 [MHz], fs=36.864[MHz], fs=32.768 [MHz] or fs=28.672 [MHz].

In a preferred embodiment, the demodulation digital circuit includes: aquadrature demodulator operable to quadrature-demodulate the signalundersampled by the sampler; and a low-pass filter operable tolow-pass-filter the signal quadrature-demodulated by the quadraturedemodulator; and a received data reproducing section operable toreproduce received data from the signal low-pass-filtered by thelow-pass filter; and the quadrature demodulator converts the signalundersampled by the sampler to a signal including a component whosecenter frequency is zero.

In a preferred embodiment, the demodulation digital circuit includes: acomplex filter operable to filter, by using a digital filter, either oneof a positive frequency component and a negative frequency component ofthe signal under sampled by the sampler whose center frequency is closerto zero; and a received data reproducing section operable to reproducereceived data from the signal filtered by the complex filter.

A fourth aspect of the present invention is directed to a wirelessdigital receiver in a wireless communications system fortransmitting/receiving a first wireless signal from a first wirelesscommunications device and a second wireless signal from a secondwireless communications device, the first and second wireless signalshaving different frequency bands from eachother, the wireless digitalreceiver receiving the second wireless signal in the first wirelesscommunications device and digitally demodulating the second wirelesssignal, the wireless digital receiver including: a frequency converteroperable to downconvert the second wireless signal transmitted from thesecond wireless communications device to a low-frequency signal whosecenter frequency is fj [Hz]; a sampler operable to oversample thelow-frequency signal downconverted by the frequency converter; and ademodulation digital circuit operable to demodulate the signaloversampled by the sampler after correcting a center frequency thereofto fi [Hz], wherein: a sampling frequency used in the sampler and thatused in the second wireless communications device are the same samplingfrequency fs [Hz]; the sampling frequency fs [Hz] is set to a value thatis an even-number multiple of a wireless symbol transmission rate suchthat oversampling is done in the sampler and undersampling is done in asampler of the second wireless communications device; and the centerfrequency fi [Hz] is ½ to 1 times a frequency corresponding to abandwidth of the first and second wireless signals and is ½^(N) (N is anatural number) times the sampling frequency fs [Hz].

In a preferred embodiment, where the bandwidth of the first and secondwireless signals is 2×Bch [Hz] and the wireless symbol transmission rateis fsym [Hz], the sampling frequency fs [Hz] and the frequency fi [Hz]are expressed as shown in the following expressions: $\begin{matrix}{{fi} = \frac{2{kfsym}}{2^{N}}} \\{{fs} = {2^{N}{fi}}}\end{matrix}$where k is an integer satisfying $\begin{matrix}{{\frac{{fd} + {Bch}}{\left( {n + 1} \right){fsym}} \leqq k \leqq \frac{{fd} - {Bch}}{n\quad{fsym}}}{and}} & {{Exp}.\quad 12} \\{k \leqq \frac{fd}{2{fsym}}} & {{Exp}.\quad 14}\end{matrix}$and N is an integer satisfying $\begin{matrix}{{\log\quad 2\left\{ \frac{{fd} + {Bch}}{\left( {n + 1} \right){Bch}} \right\}} \leqq N \leqq {\log\quad 2\left\{ \frac{2\left( {{fd} - {Bch}} \right)}{n\quad{Bch}} \right\}}} & {{Exp}.\quad 22}\end{matrix}$where n is an integer satisfying $\begin{matrix}{1 \leqq n \leqq {\frac{{fd} - {Bch}}{2{Bch}}.}} & {{Exp}.\quad 7}\end{matrix}$In a preferred embodiment, the demodulation digital circuit includes: aquadrature demodulator operable to quadrature-demodulate the signaloversampled by the sampler; an automatic frequency controller operableto correct the signal quadrature-demodulated by the quadraturedemodulator to a signal having a component whose frequency is fi [Hz]; alow-pass filter operable to low-pass-filter the signalfrequency-corrected by the automatic frequency controller; and areceived data reproducing section operable to reproduce received datafrom the signal low-pass-filtered by the low-pass filter.

In a preferred embodiment, the frequency fj [Hz] is 3.000 [MHz].

Each functional block of the present invention is preferably implementedin the form of an integrated circuit. The integrated circuitsimplementing these functional blocks may be individually formed into aseparate chip, or some or all of them may be formed together into asingle chip. In the present specification, “an integrated circuit”refers not only to an integrated circuit provided in the form of asingle chip, but also to a group of integrated circuits that are formedtogether into a single chip.

The effects of the present invention will now be described. In awireless communications system of the present invention and a wirelessdigital receiver for use therein, the sampling frequency used in thefirst wireless communications device (base station) is the same as thatused in the second wireless communications device (mobile station).Therefore, the same demodulation digital circuit for performing adigital demodulation operation can be used for the first and secondwireless communications devices. Therefore, with the present invention,it is not necessary to provide a separate demodulation digital circuitfor the first and second wireless communications devices (the basestation and the mobile station), whereby it is possible to provide aninexpensive wireless digital receiver, thereby reducing the overall costof the wireless communications system.

Moreover, with the provision of the automatic frequency controller inthe demodulation digital circuit, the local oscillator is allowed acertain degree of freedom, which also contributes to reducing the cost.

These and other objects, features, aspects and advantages of the presentinvention will become more apparent from the following detaileddescription of the present invention when taken in conjunction with theaccompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram showing a functional configuration of awireless communications system 1 according to a first embodiment of thepresent invention;

FIG. 2 is a block diagram showing a functional configuration of a firstwireless digital receiver 21;

FIG. 3 is a diagram showing pass-band characteristics of a low-passfilter 103;

FIG. 4 is a block diagram showing a functional configuration of a secondwireless digital receiver 31;

FIG. 5 is a diagram showing a configuration of a quadrature modulator;

FIG. 6A and FIG. 6B are diagrams showing the spectrum of a transmittedsignal, and the result of a multiplication of the transmitted signalwith a sine wave;

FIG. 7A and FIG. 7B are diagrams showing the spectrum of a transmittedsignal, and the result of a multiplication of the transmitted signalwith a sine wave;

FIG. 8 is a diagram showing the spectrum of a sampled signal S1(mTs)outputted from a sampler 101;

FIG. 9 is a diagram showing a configuration of a quadrature demodulator;

FIG. 10 is a diagram showing the spectrum of a sampled signal S2(mTs)obtained by sampling a modulated low-frequency signal L2(t) whose centerfrequency is fd=40.000 [MHz] with the sampling frequency fs=24.576[MHz];

FIG. 11 is a diagram showing the spectrum of the sampled signal S1(mTs)outputted from the sampler 101 where the center frequency of a modulatedlow-frequency signal L1(t) is fi=3.072 [MHz] and the sampling frequencyfs is set to 12.288 [MHz];

FIG. 12 is a diagram showing the spectrum of the sampled signal S2(mTs)obtained by sampling the modulated low-frequency signal L2(t) whosecenter frequency is fd=40.000 [MHz] with the sampling frequencyfs=12.288 [MHz];

FIG. 13 is a block diagram showing a functional configuration of thefirst wireless digital receiver 21 according to a second embodiment ofthe present invention;

FIG. 14 is a diagram showing exemplary pass-band characteristics of acomplex filter 602;

FIG. 15A, FIG. 15B, FIG. 15C, FIG. 15D, FIG. 15E and FIG. 15F arediagrams used for illustrating the pass-band characteristics of thecomplex filter 602;

FIG. 16A and FIG. 16B are block diagrams each showing a functionalconfiguration of the first wireless digital receiver 21 according to athird embodiment of the present invention;

FIG. 17 is a diagram showing the spectrum of an in-phase componentsampled signal I(mTs) and a quadrature component sampled signal Q(mTs)outputted from a quadrature demodulator 802;

FIG. 18 is a diagram showing a configuration of a base-station wirelesscommunications device 12 according to a fourth embodiment of the presentinvention;

FIG. 19 is a diagram showing a configuration of a mobile-stationwireless communications device 11 according to the fourth embodiment ofthe present invention;

FIG. 20 is a diagram schematically showing a conventional base station9000 and a conventional mobile station 9001 communicating with eachotherusing the DSRC system;

FIG. 21 is a diagram showing a general configuration of a conventionalbase-station wireless communications device employing the LOW-IFarchitecture; and

FIG. 22 is a diagram showing a general configuration of a conventionalmobile-station wireless communications device employing thesingle-conversion architecture.

BEST MODE FOR CARRYING OUT THE INVENTION First Embodiment

FIG. 1 is a block diagram showing a functional configuration of awireless communications system 1 according to a first embodiment of thepresent invention. In FIG. 1, the wireless communications system 1includes a base station 2 being a first wireless communications device,and a mobile station 3 being a second wireless communications device.The base station 2 includes a first wireless digital receiver 21 and afirst wireless transmitter 22. The mobile station 3 includes a secondwireless digital receiver 31 and a second wireless transmitter 32. FIG.1 only shows one base station 2 and one mobile station 3 for the sake ofsimplicity. In practice, however, there are a plurality of base stations2 and a plurality of mobile stations 3 communicating with one anotherusing different channels.

The wireless communications system 1 employs the DSRC system standard.Therefore, one of 5815 [MHz], 5820 [MHz], 5825 [MHz], 5830 MHz], 5835[MHz], 5840 [MHz] and 5845 [MHz], is used for the uplink from the mobilestation 3 to the base station 2. Herein, the center frequency of asignal used for the uplink is denoted as fc [Hz] (hereinafter referredto simply as “fc”).

For the downlink from the base station 2 to the mobile station 3, one of5775 [MHz], 5780 [MHz], 5785 [MHz], 5790 [MHz], 5795 [MHz], 5800 [MHz]and 5805 [MHz] is used according to the frequency fc for the uplink. Thefrequency fd [Hz] (hereinafter referred to simply as “fd”), which isequal to the difference between the frequency used for the downlink andthat used for the uplink, is always 40.000 [MHz]. The center frequencyof a signal used for the downlink is fc-fd [Hz] (hereinafter referred tosimply as “fc-fd”).

The DSRC system standard specifications include a section on thetechnical requirements for wireless equipment, which specifies variousrequirements for the base station and the mobile station.

The DSRC system standard specifies that the bandwidth of a signal ineach channel (hereinafter referred to as the “channel bandwidth”) is 5[MHz]. According to this specification, it is understood that where thechannel bandwidth is denoted as 2×Bch, Bch=2.5 [MHz].

In the DSRC system, it is specified that the modulation scheme employedneeds to be either the ASK (Amplitude Shift Keying) scheme in which thewireless symbol frequency fsym [Hz] (hereinafter referred to simply as“fsym”) is 1.024 [MHz] or the π/4 shift QPSK (Quadrature Phase ShiftKeying) in which the wireless symbol frequency fsym is 2.048 [MHz]. TheDSRC system of the present embodiment employs the π/4 shift QPSK schemewith the wireless symbol frequency fsym=2.048 [MHz]. In the case of theASK scheme, since the Manchester coding is used, fsym=1.024 [MHz]converted to a baud rate is 2.048 [MHz], which is equal to that of theπ/4 shift QPSK scheme. Therefore, the following description in principleapplies to the ASK scheme.

The first wireless transmitter 22 of the base station 2 outputs a signal(first wireless signal) whose center frequency is fc-fd. In response,the second wireless digital receiver 31 of the mobile station 3 receivesthe signal (first wireless signal) whose center frequency is fc-fd. Thesecond wireless digital receiver 31 downconverts the received signal(first wireless signal) whose center frequency is fc-fd to a signalwhose center frequency is fd=40.000 [MHz]. The second wireless digitalreceiver 31 undersamples the signal whose center frequency is fd insynchronism with the sampling signal whose sampling frequency isfs=24.576 [MHz]. The second wireless digital receiver 31 demodulates theundersampled signal by using a digital circuit to obtain received data.

The second wireless transmitter 32 of the mobile station 3 outputs asignal (second wireless signal) whose center frequency is fc. Inresponse, the first wireless digital receiver 21 of the base station 2receives the signal second wireless signal) whose center frequency isfc. The first wireless digital receiver 21 downconverts the receivedsignal (second wireless signal) whose center frequency is fc to a signalwhose center frequency is fi=3.072 [MHz]. The first wireless digitalreceiver 21 oversamples the signal whose center frequency is fi insynchronism with the sampling signal whose sampling frequency isfs=24.576 [MHz]. The first wireless digital receiver 21 demodulates theoversampled signal by using a digital circuit to obtain received data.

FIG. 2 is a block diagram showing a functional configuration of thefirst wireless digital receiver 21. In FIG. 2, the first wirelessdigital receiver 21 includes a frequency converter 100, a sampler 101, aquadrature demodulator 102, a low-pass filter 103, a sampling signalgenerator 104 and a received data reproducing section 105. Thequadrature demodulator 102, the low-pass filter 103 and the receiveddata reproducing section 105 will be hereinafter referred tocollectively as a “first demodulation digital circuit”. Assume that amodulated high-frequency signal R1(t) whose center frequency is fcisinputted to the first wireless digital receiver 21.

The frequency converter 100 downconverts the modulated high-frequencysignal R1(t) and outputs the modulated low-frequency signal L1(t) whosecenter frequency is fi=3.072 [MHz] The reason why the signal isdownconverted to fi=3.072 [MHz] will later be described in detail.

The sampling signal generator 104 outputs a sampling signal whosesampling frequency is fs=24.576 [MHz]. The reason why the samplingfrequency is fs=24.576 [MHz] will later be described in detail.

The sampler 101 oversamples the modulated low-frequency signal L1(t) insynchronism with the sampling signal outputted from the sampling signalgenerator 104 to output a sampled signal S1(mTs). Herein, m is aninteger (m=. . . , −1, 0, 1, . . . ), and Ts is the sampling period,i.e., Ts=1/fs.

The quadrature demodulator 102 performs an operation of multiplying thesampled signal S1(mTs) outputted from the sampler 101 by exp(−jθ×mTs)(where j is the imaginary unit) to output two signals whose phases aredifferent from each other by π/2 [rad], i.e., an in-phase componentsampled signal I1(mTs) and a quadrature component sampled signalQ1(mTs). Herein, θ is set to a value such that a signal that isfrequency-shifted so that the center frequency thereof is zero isincluded in the signals outputted from the quadrature demodulator 102after the multiplication with exp(−jθ×mTs). The value 8 will later bedescribed in detail.

FIG. 3 is a diagram showing pass-band characteristics of the low-passfilter 103. A low-pass filter 203 is a digital filter whose frequencypass band is zero to Bch/2. With the provision of the low-pass filter103, the in-phase component Ib1(mTs) and the quadrature componentQb1(mTs) of the baseband quadrature demodulated signal outputted fromthe low-pass filter 103 will only have a frequency component that isfrequency shifted so that the center frequency thereof is zero.

The signals Ib1(mTs) and Qb1(mTs) outputted from the low-pass filter 103only have a component that is frequency-shifted so that the centerfrequency thereof is zero, whereby the received data reproducing section105 can output the received data by means of delay detection, or thelike.

FIG. 4 is a block diagram showing a functional configuration of thesecond wireless digital receiver 31. In FIG. 4, the second wirelessdigital receiver 31 includes a frequency converter 200, a sampler 201, aquadrature demodulator 202, the low-pass filter 203, a sampling signalgenerator 204 and a received data reproducing section 205. Thequadrature demodulator 202, the low-pass filter 203 and the receiveddata reproducing section 205 will be hereinafter referred tocollectively as a “second demodulation digital circuit”. Assume that amodulated high-frequency signal R2(t) whose center frequency is fc-fd isinputted to the second wireless digital receiver 31.

The frequency converter 200 downconverts the modulated high-frequencysignal R2(t) to output the modulated low-frequency signal L2(t) whosecenter frequency is fd=40.000 [MHz]. In the mobile station of the DSRCsystem, a local oscillator signal whose frequency is fc is outputtedfrom a local oscillator (not shown) in order to output a signal to betransmitted. The mobile station employs a single-conversion architectureusing the local oscillator signal. Moreover, the frequency of the signalthat the mobile station receives is fc-fd. Therefore, the frequencyconverter 200 downconverts the modulated high-frequency signal R2(t) tofd=40.000 [MHz] by using the local oscillator signal whose frequency isfc.

The sampling signal generator 204 outputs a sampling signal whosesampling frequency is fs=24.576 [MHz]. Thus, the sampling signalgenerator 204 is the same as the sampling signal generator 104 in thefirst wireless digital receiver 21. As described above, in the presentembodiment, the sampling frequency used in the first wireless digitalreceiver 21 is equal to that used in the second wireless digitalreceiver 31. The reason why the same sampling frequency can be used willlater be described. Also, the reason why the sampling frequency isfs=24.576 [MHz] will later be described in detail.

The sampler 201 undersamples the modulated low-frequency signal L2(t) insynchronism with the sampling signal outputted from the sampling signalgenerator 204 to output a sampled signal S2(mTs). Herein, m is aninteger (m=. . . , −1, 0, 1, . . . ), and Ts is the sampling period,i.e., Ts=1/fs.

The quadrature demodulator 202 performs an operation of multiplying thesampled signal S2(mTs) outputted from the sampler 201 by exp(−jη×xmTs)(where j is the imaginary unit) to output two signals whose phases aredifferent from each other by π/2 [rad], i.e., an in-phase componentsampled signal I2(mTs) and a quadrature component sampled signalQ2(mTs). Herein, η is set to a value such that a signal that isfrequency-shifted so that the center frequency thereof is zero isincluded in the signals outputted from the quadrature demodulator 202after the multiplication with exp(−jη×mTs). The value η will later bedescribed in detail. As will be described later, η is a value differentfrom θ used in the quadrature demodulator 202 of the first wirelessdigital receiver 21. Thus, the quadrature demodulator 102 used in thefirst wireless digital receiver 21 is the same as the quadraturedemodulator 202 used in the second wireless digital receiver 31 exceptthat the rotation angles θ and η used in the multiplication with exp aredifferent from each other. The rotation angles being different from eachother will later be described in detail.

The low-pass filter 203 is a digital filter whose frequency pass band iszero to Bch/2, as is the low-pass filter 103 in the first wirelessdigital receiver 21. Thus, FIG. 3 is relied upon also for the low-passfilter 203. With the provision of the low-pass filter 203, the in-phasecomponent Ib2(mTs) and the quadrature component Qb2(mTs) of the basebandquadrature demodulated signal outputted from the low-pass filter 203will only have a frequency component that is frequency-shifted so thatthe center frequency thereof is zero. The low-pass filter 103 used inthe first wireless digital receiver 21 is the same as the low-passfilter 203 used in the second wireless digital receiver 31.

The signal Ib2(mTs) and Qb2(mTs) outputted from the low-pass filter 203only have a component that is frequency-shifted so that the centerfrequency thereof is zero, whereby the received data reproducing section205 can output the received data by means of delay detection, or thelike.

Since the sampling frequency for the first wireless digital receiver 21is equal to that for the second wireless digital receiver 31, the samesampling signal generator can be used as the sampling signal generators104 and 204. Moreover, the same sampler can be used as the samplers 101and 201. Furthermore, the same low-pass filter can be used as thelow-pass filters 103 and 203. In addition, since it is only necessary tochange the rotation angle between the quadrature demodulator 102 and thequadrature demodulator 202, the same quadrature demodulator can be usedas the quadrature demodulator 102 in the first wireless digital receiver21 and the quadrature demodulator 202 in the second wireless digitalreceiver 31 if two different rotation angles can be stored in a memorydevice and the quadrature demodulator used is capable of switching therotation angle values from one to another.

Now, the reason why sampling is done properly at the samplers 101 and201 and the received data can be properly obtained at the first andsecond wireless digital receivers 21 and 31 by using fi=3.072 [MHz] andfs=24.576 [MHz] will be described. Specifically the reason why thereceived data can be restored completely as the sampler 101 oversamplesa signal whose center frequency is fi=3.072 [MHz] with the samplingfrequency fs=24.576 [MHz] while the received data can be restoredcompletely as the sampler 201 undersamples a signal whose centerfrequency is fi=40.000 [MHz] with the sampling frequency fs=24.576 [MHz]will be described.

A transmitted signal can generally be expressed as shown in Expression 1below, using a complex signal.Re[S(t)exp{j(ωct+φ)}]  Exp. 1

This is because the transmitted baseband signal S(t) is, in the firstplace, a complex signal expressed as TXI+jTxQ, which isquadrature-modulated (multiplied with Expression 2) by using aquadrature modulator as shown in FIG. 5 and then outputted as a radiowave.exp{j(ωct+φ)}  Exp. 2

The receiving side downconverts the transmitted signal by multiplying itwith a sine wave. First, the transmitted signal and the sine wave can beexpressed by using a complex signal as shown in Expression 3 andExpression 4, respectively. $\begin{matrix}{{{Transmitted}{\quad\quad}{Signal}}{{{Re}\left\lbrack {{S(t)}\exp\left\{ {j\left( {{\omega\quad c\quad t} + \phi} \right)} \right\}} \right\rbrack} = {\frac{1}{2}\left\{ {{{S(t)}\exp\left\{ {j\left( {{\omega\quad c\quad t} + \phi} \right)} \right\}} + {{S^{*}(t)}\exp\left\{ {j\left( {{\omega\quad c\quad t} + \phi} \right)} \right\}^{*}}} \right\}}}} & {{Exp}.\quad 3} \\{{Sine}\quad{Wave}} & \quad \\{{\cos\left\{ {{\left( {{\omega\quad c} - {\omega\mathbb{i}}} \right)t} + \phi} \right\}} = {\frac{1}{2}\left\{ {{\exp\left\lbrack {j\left\{ {{\left( {{\omega\quad c} - {\omega\quad{\mathbb{i}}}} \right)t} + \phi} \right\}} \right\rbrack} + {\exp\left\lbrack {j\left\{ {{\left( {{\omega\quad c} - {\omega\quad{\mathbb{i}}}} \right)t} + \phi} \right\}} \right\rbrack}^{*}} \right\}}} & {{Exp}.\quad 4}\end{matrix}$

According to Expression 3, the spectrum of the transmitted signal can beexpressed on a plane as shown in FIG. 6A where the horizontal axisrepresents the complex frequency and the vertical axis represents thespectral intensity.

It can be seen from FIG. 6A that the transmitted signal is a signal madeup of a spectrum of S(t) at a center angular frequency of +ωc andanother spectrum of S*(t) at a center angular frequency of −ωc.

Similarly, it can be seen from Expression 4 that the sine wave is asignal made up of a sine-wave signal whose center angular frequency is+ωc and another sine-wave signal whose center angular frequency is −ωc.

Where the center angular frequency of the local oscillator used for thedown conversion is ωc−ωi, the frequency-converted signal obtained bymultiplying the transmitted signal by the sine wave can be expressed asshown in Expression 5. $\begin{matrix}{{{{Re}\left\lbrack {{S(t)}\exp\left\{ {j\left( {{\omega\quad c\quad t} + \phi} \right)} \right\}} \right\rbrack}\cos\left\{ {{\left( {{\omega\quad c} - {\omega\quad{\mathbb{i}}}} \right)t} + \phi} \right\}} = {\frac{1}{4}\left\lbrack {{{S(t)}\exp\left\{ {j\left( {{{\omega\mathbb{i}}\quad t} + \phi - \phi} \right)} \right\}} + {{S(t)}\exp\left\{ {j\left( {{\omega\quad{\mathbb{i}}\quad t} + \phi - \phi} \right)} \right\}^{*}}} \right\rbrack}} & {{Exp}.\quad 5}\end{matrix}$

FIG. 6B diagramatically shows the downconversion represented byExpression 5. It can be seen that when the transmitted signal isfrequency-converted so that the center angular frequency is ωi (afrequency value as close to zero as possible while the intended wavedoes not contain a DC component), an adjacent channel (ch1-) falls intothe band of the intended wave, which thus becomes a disturbing wave. Inprinciple, ch1- can be removed by using, for example, an image rejectionmixer (see p. 281 of Non-Patent Document 1). In practice, however, thesuppression can only be done by about 30 to 40 dB at best due to aquadrature error between the in-phase component and the quadraturecomponent of the quadrature demodulated signal, as known in the art (seeNon-Patent Document 2). However, since the adjacent wave selectivity at5 [MHz] intervals is specified in the DSRC system neither for the basestation nor for the mobile station (see STD-T75, Ver. 1.2, P. 33), ch1-does not have to be suppressed completely.

FIG. 7B shows the result of gradually moving away the center angularfrequency ωi in the positive direction from its position shown in FIG.6B. As shown in FIG. 7B, the next adjacent channel ch2- (the nextadjacent channel is defined as a 10 [MHz] -interval signal in STD-T75,Ver. 1.2, P. 33) falls into the band of the intended wave. In such acase, if the ch2- removal deteriorates even by a small degree, themargin from the standard 15 dB will decrease, and the standard will nolonger be met in worst cases. Therefore, it is preferred that the centerangular frequency is as close to zero as possible.

In view of the above, when employing a LOW-IF architecture for the basestation of the DSRC system, it is preferred that settings are made sothat the adjacent channel ch1-* falls into the intended wave-band. Thus,where the center frequency of the down converted signal is fi,Expression 6 below should be satisfied.Bch≦fi≦2Bch   Exp. 6where fi=2πωi and 2Bch is bandwidth per channel

Since 2×Bch=5 [MHz] and fd=40.000 [MHz], a comparison between fi and fdyields fi<fd. Therefore, when a signal whose center frequency is fi andanother signal whose center frequency is fd are to be sampled with thesame sampling frequency, the signal whose center frequency is fi will beoversampled while the signal whose center frequency is fd will beundersampled. A sampling frequency such that the signals will both beoversampled can be used. In such a case, however, the sampling frequencywill be very high, and it will be necessary to use a sampler capable ofhandling high-frequency signals, thereby making it difficult to realizethe circuit at a low cost.

Therefore, the signal to be undersampled is a signal whose centerfrequency is fd. A necessary and sufficient condition for realizing theundersampling operation is Expression 7 and Expression 8 below (seeNon-Patent Document 3, p. 123, Expressions B.12 and B.16).$\begin{matrix}{1 \leqq n \leqq \frac{{fd} - {Bch}}{2{Bch}}} & {{Exp}.\quad 7} \\{\frac{2\left( {{fd} + {Bch}} \right)}{n + 1} \leqq {fs} \leqq \frac{2\left( {{fd} - {Bch}} \right)}{n}} & {{Exp}.\quad 8}\end{matrix}$Herein, fs represents the sampling frequency.

The signal to be oversampled is a signal whose center frequency is fi.According to the Nyquist's theorem, a necessary and sufficient conditionfor realizing the oversampling operation is Expression 9 below.fs≧2Bch   Exp. 9

Moreover, the condition for easily realizing the demodulation digitalcircuit is generally represented by Expression 10 below.fs=2^(N)fi=2kfsym   Exp. 10Herein, N and k are integers, and fsym is a frequency representing thewireless symbol transmission rate.

Expression 8 and Expression 10 yield Expression 11. $\begin{matrix}{\frac{2\left( {{fd} + {Bch}} \right)}{n + 1} \leqq {2{kfsym}} \leqq \frac{2\left( {{fd} - {Bch}} \right)}{n}} & {{Exp}.\quad 11}\end{matrix}$

Expression 11 can be rearranged with respect to k, yielding Expression12. $\begin{matrix}{\frac{{fd} + {Bch}}{\left( {n + 1} \right){fsym}} \leqq k \leqq \frac{{fd} - {Bch}}{n\quad{fsym}}} & {{Exp}.\quad 12}\end{matrix}$

The conditions for k will be further discussed. As described above, theundersampling scheme is used for a signal whose center frequency is fd.Considering this fact together with Expression 10 yields Expression 13.fd≧fs=2kfsym   Exp. 13

This can be transformed with respect to k, yielding Expression 4.$\begin{matrix}{k \leqq \frac{fd}{2{fsym}}} & {{Exp}.\quad 14}\end{matrix}$

Note that since Expression 15 holds true except when n=1 (i.e., it holdstrue when n≧2), Expression 14 is an expression that always holds true aslong as Expression 11 holds true. $\begin{matrix}{{fd} > \frac{2\left( {{fd} - {Bch}} \right)}{n}} & {{Exp}.\quad 15}\end{matrix}$

Rearranging Expression 6 by using Expression 10 yields Expression 16.2^(N)Bch≦fs≦2^(N+1)Bch   Exp. 16

Next, the condition under which Expression 8 and Expression 16 aresatisfied at the same time will be considered. First, the conditionunder which Expression 8 and Expression 16 are not satisfied at the sametime will be considered. The condition under which there is no solutionthat satisfies Expression 8 and Expression 16 at the same time is asshown in Expression 17. $\begin{matrix}{{2^{N + 1}{Bch}} < {\frac{2\left( {{fd} + {Bch}} \right)}{n + 1}{\quad\quad}{or}\quad 2^{N}{Bch}} > \frac{2\left( {{fd} - {Bch}} \right)}{n}} & {{Exp}.\quad 17}\end{matrix}$

Rearranging Expression 17 with respect to N yields Expression 18.$\begin{matrix}{N < {\log\quad 2\left\{ \frac{{fd} + {Bch}}{\left( {n + 1} \right){Bch}} \right\}\quad{or}\quad N} > {\log\quad 2\left\{ \frac{2\left( {{fd} - {Bch}} \right)}{n\quad{Bch}} \right\}}} & {{Exp}.\quad 18}\end{matrix}$

Now, a calculation as shown in Expression 19 below is done for acomparison between antilogarithms in Expression 18. $\begin{matrix}\begin{matrix}{{\frac{2\left( {{fd} - {Bch}} \right)}{n\quad{Bch}} - \frac{{fd} + {Bch}}{\left( {n + 1} \right){Bch}}} = \frac{{\left( {n + 1} \right){fd}} - {\left( {{3n} + 1} \right){Bch}}}{{n\left( {n + 1} \right)}{Bch}}} \\{= \frac{{fd} - {\left( {3 - \frac{2}{n + 1}} \right){Bch}}}{n\quad{Bch}}}\end{matrix} & {{Exp}.\quad 19}\end{matrix}$

Since 1≦n based on Expression 7, 2/(n+1)≦1. Thus, Expression 20 can beobtained from Expression 19. $\begin{matrix}{\frac{{fd} - {3{Bch}}}{n\quad{Bch}} < \frac{{fd} - {\left( {3 - \frac{2}{n + 1}} \right){Bch}}}{n\quad{Bch}} \leqq \frac{{fd} - {2{Bch}}}{n\quad{Bch}}} & {{Exp}.\quad 20}\end{matrix}$

Since fd=40.000 [MHz]=16Bch in the DSRC system, the value of Expression20 is greater than zero. Thus, Expression 21 holds true. $\begin{matrix}{\frac{2\left( {{fd} - {Bch}} \right)}{n\quad{Bch}} > \frac{{fd} + {Bch}}{\left( {n + 1} \right){Bch}}} & {{Exp}.\quad 21}\end{matrix}$

Therefore, the condition for satisfying Expression 8 and Expression 16at the same time can be obtained by negating Expression 18 as shown inExpression 22. $\begin{matrix}{{\log\quad 2\left\{ \frac{{fd} + {Bch}}{\left( {n + 1} \right){Bch}} \right\}} \leqq N \leqq {\log\quad 2\left\{ \frac{2\left( {{fd} - {Bch}} \right)}{n\quad{Bch}} \right\}}} & {{Exp}.\quad 22}\end{matrix}$

Thus, values of fi and fs of the present invention are obtained by firstobtaining a value of n that satisfies Expression 7. Then, the value of kthat satisfies Expression 12 and Expression 14 is obtained. Then, forthe obtained value of n, the value of N that satisfies Expression 22 isobtained. Then, the obtained values of N and k are substituted intoExpression 10 to obtain fi. Then, the obtained values of N and fi aresubstituted into Expression 10 to obtain fs.

Now, values of fi and fs will be obtained with an actual DSRC system. Inthe DSRC system, it is presumed that Bch=2.5 [MHz], fd=40.000 [MHz] andfsym=2.048 [MHz].

First, an integer that satisfies Expression 7 is derived. In theillustrated example, n=1, 2, . . . , 7 satisfies Expression 7.

Then, one of the integer values of n is selected, and an integer k thatsatisfies Expression 12 and Expression 14 is derived. With a certaininteger n (1≦n≦7), there may possibly be no integer k that satisfiesExpression 12 and Expression 14. Specifically, when n=1, 4, 5 and 7,there is no integer k that satisfies Expression 12 and Expression 14.When n=2, k=7, 8 or 9. When n=3, k=6. When n=6, k=3.

Then, one of the integer values n (1≦n≦7) satisfying Expression 7 isselected, and an integer N that satisfies Expression 22 is derived. Whenn=1, N=4. When n=2 or 3, N=3. When n=4, 5, 6 or 7, N=2.

Finally, based on Expression 10, the value fi for the integers k and Nis obtained, based on which the value fs is obtained.

Table 1 below shows possible combinations of n, k and N, and the valuesfi and fs therefor. TABLE 1 n N k fi[MHz] fs[MHz] 1 4 — — — 2 3 9 4.60836.864 2 3 7 3.584 28.672 2 3 8 4.096 32.768 3 3 6 3.072 24.576 4 2 — —— 5 2 — — — 6 2 3 3.072 12.288 7 2 — — —

In Table 1, “-” means that there is no value that satisfies theconditions described above.

As can be seen from Table 1, the minimum value for fi is 3.072 [MHz],and as can be seen from the description above with reference to FIG. 6A,FIG. 6B, FIG. 7A and FIG. 7B, the falling of the next adjacent channelch2- into the intended wave band is least significant when fi is 3.072[MHz]. Thus, the following description will be limited to a case wherefi is 3.072 [MHz]. When fi is 3.072 [MHz], fs is 24.576 [MHz] or 12.288[MHz] based on Table 1. In the present embodiment, 24.576 [MHz] is usedas fs.

The description above shows that the received data can be restoredcompletely as the sampler 101 oversamples a signal whose centerfrequency is fi=3.072 [MHz] with the sampling frequency fs=24.576 [MHz]while the received data can be restored completely as the sampler 201undersamples a signal whose center frequency is fi=40.000 [MHz] with thesampling frequency fs=24.576 [MHz] While fi=3.072 [MHz] and fs=24.576[MHz] in the illustrated example, a combination of fi=3.072 [MHz] andfs=12.288 [MHz] may also be used as can be seen from Table 1. Moreover,as can be seen from Table 1, other possible combinations include:fi=4.608 [MHz] and fs=36.864 [MHz]; fi=4.096 [MHz] and fs=36.768 [MHz];and fi=3.584 [MHz] and fs=28.672 [MHz]. Note that fd=40.000 [MHz] in anycase.

FIG. 8 is a diagram showing the spectrum of the sampled signal S1(mTs)outputted from the sampler 101. In FIG. 8, the horizontal axisrepresents the complex frequency and the vertical axis represents thepower spectral intensity.

In FIG. 8, 2Bch represents the channel bandwidth, and 2Bch=5 [MHz] inthe DSRC system. In FIG. 8, a spectrum 300 represents the spectrum ofthe modulated low-frequency signal L1(t). The other spectra are foldingspectra occurring as a result of sampling the modulated low-frequencysignal L1(t) with a sampling period of Ts. The figure shows, as foldingspectra, a signal whose center frequency is fs±fi and another signalwhose center frequency is −fs±fi.

The quadrature demodulator 102 receives the sampled signal S1(mTs)outputted from the sampler 101, and outputs two signals whose phases aredifferent from each other by π/2 [rad], i.e., the in-phase componentsampled signal I1(mTs) and the quadrature component sampled signalQ1(mTs). Specifically, the quadrature demodulator 102 performs anoperation S1(mTs)×exp(−jθ×mTs) by using θ[rad] expressed as shown inExpression 23 to obtain the in-phase component sampled signal I1(mTs)and the quadrature component sampled signal Q1(mTs). $\begin{matrix}{\theta = {\frac{1}{2^{N - 1}}\pi}} & {{Exp}.\quad 23}\end{matrix}$

Herein, N is as shown in Table 1. Specifically, where fi=3.072 [MHz] andfs=24.576 [MHz], N=3. Where fi=3.072 [MHz] and fs=12.288 [MHz], N=2.Where fi=4.608 [MHz] and fs=36.864 [MHz], N=3.

For example, a signal Sb(mTs) whose center frequency is fb [Hz] beingnot zero and which has been sampled with the sampling frequency fs canbe converted to a signal whose center frequency is zero using a digitalcircuit by multiplying Sb(mTs) by exp(−j2π×fb/fs×t) and shifting thefrequency by fb in the positive direction. While t represents the time,inside a digital circuit whose sampling frequency is fs, t cannot takecontinuous values but takes discrete values at regular intervals of Ts.Therefore, for every Ts, the value by which Sb(mTs) should be multipliedis expressed as shown in Expression 24. $\begin{matrix}{\exp\left( {{- j}\quad 2\quad\pi\frac{fb}{fs}m} \right)} & {{Exp}.\quad 24}\end{matrix}$

Thus, Sb(mTs) whose center frequency is fb can be converted to a signalwhose center frequency is zero by performing an operation as shown inExpression 25. $\begin{matrix}{{{Sb}({mTs})} \times {\exp\left( {{- j}\quad 2\quad\pi\frac{fb}{fs}m} \right)}} & {{Exp}.\quad 25}\end{matrix}$

Expanding the exp term using the Euler's formula yields Expression 26.$\begin{matrix}{{\exp\left( {{- j}\quad 2\quad\pi\frac{fb}{fs}m} \right)} = {{\cos\left\{ {\left( {2\quad\pi\frac{fb}{fs}} \right)m} \right\}} - {j\quad\sin\left\{ {\left( {2\quad\pi\frac{fb}{fs}} \right)m} \right\}}}} & {{Exp}.\quad 26}\end{matrix}$

Thus, the operation of Expression 25can be realized by using a circuitconfiguration as shown in FIG. 9. FIG. 9 is a schematic diagram showinga configuration of a circuit for shifting the frequency by -fb to obtaina signal whose center frequency is zero. As shown in FIG. 9, the circuitoutputs Ib(mTs) being an in-phase component of Sb(mTs) and Qb(mTs) beinga quadrature component thereof. Therefore, the circuit shown in FIG. 9can be considered a quadrature demodulator. Thus, FIG. 9 shows aninternal configuration of the quadrature demodulators 102 and 202.

Therefore, θ in the operation S1(mTs)×exp(−jθ×m) performed by thequadrature demodulator 102 is determined as shown in Expression 27 belowbased on Expression 10. $\begin{matrix}{\theta = {{2\quad\pi\frac{fb}{fs}} = {{2\quad\pi\frac{fi}{fs}} = {{2\quad\pi\frac{fi}{2^{N}{fi}}} = {\frac{1}{2^{N - 1}}\pi}}}}} & {{Exp}.\quad 27}\end{matrix}$It can be seen that Expression 27 is equal to Expression 23.

Thus, as a result of the operation S1(mTs)×exp(−jθ×mTs) performed by thequadrature demodulator 102, the in-phase component sampled signalI1(mTs) and the quadrature component sampled signal Q1(mTs) each have afrequency component that is frequency-shifted so that the centerfrequency of the spectrum 300 shown in FIG. 8 is zero.

FIG. 10 is a diagram showing the spectrum of the sampled signal S2(mTs)obtained by sampling the modulated low-frequency signal L2(t) whosecenter frequency is fd=40.000 [MHz] with the sampling frequencyfs=24.576 [MHz]. In FIG. 10, the horizontal axis represents the complexfrequency. The vertical axis represents the power spectral intensity.

In FIG. 10, a spectrum 500 represents the spectrum of the modulatedlow-frequency signal L2(t), and the other spectra are folding spectraoccurring as a result of sampling the modulated low-frequency signalL2(t) with a sampling period of Ts. Spectra 501, 502 and 503 are eachspaced away from the spectrum 500 by a distance of an integer multipleof the sampling frequency, and thus the spectra 500, 501, 502 and 503are signals equivalent to one another.

However, the other spectra are not spaced apart from the spectrum 500representing the modulated low-frequency signal L2(t) by an integermultiple of the sampling frequency, and thus are spectra of signals eachhaving a frequency component different from the spectrum 500.

As described above, the spectrum 500 is the same as the spectrum 502.The quadrature demodulator 202 performs an operation S2(mTs)×exp(−jη×m)using η [rad] expressed as shown in Expression 28 to obtain the in-phasecomponent sampled signal I2(mTs) and the quadrature component sampledsignal Q2(mTs) each having a frequency component that isfrequency-shifted so that the center frequency of the spectrum 502 shownin FIG. 10 is zero. $\begin{matrix}{\eta = {{- \frac{{M\quad{fs}} - {fd}}{fs}}2\quad\pi}} & {{Exp}.\quad 28}\end{matrix}$

The basis for Expression 28 will now be described. The signal S2(mTs)outputted from the sampler 201 is equivalent to the spectrum 500, asshownin FIG. 10, and includes a signal whose center frequency is closestto zero. The center frequency of this signal can be expressed as −Mfs+fdusing a positive integer M. In FIG. 10, M=2. It can be seen fromExpression 29 that a signal whose center frequency is −Mfs+fd can befrequency-shifted to a signal whose center frequency is zero by using ηin Expression 28 above. $\begin{matrix}{\eta = {{2\quad\pi\frac{fb}{fs}} = {{2\quad\pi\frac{{{- M}\quad{fs}} + {fd}}{fs}} = {{- \frac{{M\quad{fs}} - {fd}}{fs}}2\quad\pi}}}} & {{Exp}.\quad 29}\end{matrix}$

As described above, also where fi=3.072 [MHz] and fs=12.288 [MHz], thereceived data can be properly obtained by performing a similaroperation. FIG. 11 is a diagram showing the spectrum of the sampledsignal S1(mTs) outputted from the sampler 101 in a case where the centerfrequency of the modulated low-frequency signal L1(t) is fi=3.072 [MHz]and the sampling frequency fs is 12.288 [MHz]. In FIG. 11, thehorizontal axis represents the complex frequency and the vertical axisrepresents the power spectral intensity.

In FIG. 11, 2Bch represents the channel bandwidth, and 2Bch=5 [MHz] inthe DSRC system. In FIG. 11, a spectrum 400 represents the spectrum ofthe modulated low-frequency signal L1(t) . The other spectra are foldingspectra occurring as a result of sampling the modulated low-frequencysignal L1(t) with a sampling period of Ts. The figure shows, as foldingspectra, a signal whose center frequency is fs±fi and another signalwhose center frequency is −fs±fi.

The quadrature demodulator 102 receives the sampled signal S1(mTs)outputted from the sampler 101, and outputs two signals whose phases aredifferent from each other by π/2 [rad], i.e., the in-phase componentsampled signal I1(mTs) and the quadrature component sampled signalQ1(mTs). The quadrature demodulator 102 can be configured to perform anoperation S1(mTs) ×exp(×jθ×m) using θ [rad] expressed as shown inExpression 23, as described above, to obtain the in-phase componentsampled signal I1(mTs) and the quadrature component sampled signalQ1(mTs). Also in such a case, by using the low-pass filter 103 havingpass-band characteristics as shown in FIG. 3, it is possible to obtainIb1(mTs) being an in-phase component signal of the baseband quadraturedemodulated signal and Qb1(mTs) being a quadrature component signalthereof each having only a frequency component that is frequency-shiftedso that the center frequency of the spectrum 400 is zero. Thus, thereceived data can be obtained by the received data reproducing section105.

FIG. 12 is a diagram showing the spectrum of the sampled signal S2(mTs)obtained by sampling the modulated low-frequency signal L2(t) whosecenter frequency is fd=40.000 [MHz] with the sampling frequencyfs=12.288 [MHz]. In FIG. 12, the horizontal axis represents the complexfrequency. The vertical axis represents the power spectral intensity.

In FIG. 12, a spectrum 704 represents the spectrum of the modulatedlow-frequency signal L2(t), and the other spectra are folding spectraoccurring as a result of sampling the modulated low-frequency signalL2(t) with a sampling period of Ts. Spectra 705, 706 and 707 are eachspaced away from the spectrum 704 by a distance of an integer multipleof the sampling frequency, and thus the spectra 704, 705, 706 and 707are signals equivalent to one another.

However, the other spectra are not spaced apart from the spectrum 704representing the modulated low-frequency signal L2(t) by an integermultiple of the sampling frequency, and thus are spectra of signals eachhaving a frequency component different from the spectrum 704.

As described above, the spectrum 704 is the same as the spectrum 707.The quadrature demodulator 202 performs an operation S2(mTs) ×exp(−jη×m)using η[rad] expressed as shown in Expression 28 to obtain the in-phasecomponent sampled signal I2(mTs) and the quadrature component sampledsignal Q2(mTs) each having a frequency component that isfrequency-shifted so that the center frequency of the spectrum 707 shownin FIG. 12 is zero. In this case, since the spectrum 707 isfrequency-shifted so that the center frequency thereof is zero, M=3 inExpression 28.

Also where fi=4.608 [MHz] and fs=36.864 [MHz], the received data can beproperly obtained at the first and second wireless digital receivers 21and 31 similarly by determining the rotation angles θ and η at thequadrature demodulators 102 and 202 and performing the quadraturedemodulation.

Thus, in the first embodiment, the sampling frequency in the basestation and that in the mobile station are both set to the same value fs[Hz] being an even-number multiple of the wireless symbol transmissionrate such that oversampling is done in the base station whileundersampling is done in the mobile station. Moreover, the centerfrequency fi [Hz] of a signal that has been downconverted in the basestation is ½ to 1 times the frequency corresponding to the bandwidth ofthe transmitted/received wireless signal and is ½^(N) (N is a naturalnumber) times the sampling frequency. For example, fi=3.072 [MHz] andfs=24.576 [MHz]. Alternatively, fi=3.072 [MHz] and fs=12.288 [MHz].Alternatively, fi=4.608 [MHz] and fs=36.864 [MHz], fi=3.584 [MHz] andfs=28.672 [MHz], or fi=4.096 [MHz] and fs=32.768 [MHz]. Thus, thedemodulation digital circuit in the base station and that in the mobilestation can be the same except for using different rotation angles inthe quadrature demodulation. Therefore, it is possible to providewireless digital receivers for the base station and for the mobilestation at a low cost while reducing the overall cost of the wirelesscommunications system.

Note that in FIG. 2 being a block diagram showing a functionalconfiguration of the first wireless digital receiver 21, the frequencyconverter 100, the sampler 101, the quadrature demodulator 102, thelow-pass filter 103, the sampling signal generator 104 and the receiveddata reproducing section 105 are typically each implemented in the formof an LSI being an integrated circuit. These components may beindividually formed into a separate chip, or some or all of them may beformed together into a single chip.

Moreover, in FIG. 4 being a block diagram showing a functionalconfiguration of the second wireless digital receiver 31, the frequencyconverter 200, the sampler 201, the quadrature demodulator 202, thelow-pass filter 203, the sampling signal generator 204 and the receiveddata reproducing section 205 are typically each implemented in the formof an LSI being an integrated circuit. These components may beindividually formed into a separate chip, or some or all of them may beformed together into a single chip.

While the term “LSI” is used herein as the type of integrated circuitused in the present invention, integrated circuits are also called“ICs”, “system LSIs”, “super LSIs” or “ultra LSIs” depending on thedegree of integration. Moreover, the form of an integrated circuit thatcan be used with the present invention is not limited to an LSI, but mayalternatively be a dedicated circuit or a general-purpose processor. Itmay alternatively be an FPGA (Field Programmable Gate Array) beingprogrammable after the LSI is manufactured, or a reconfigurableprocessor in which the interconnections and settings of circuit cells inthe LSI can be reconfigured. Furthermore, if advancements in thesemiconductor technology or derivative technologies bring forth a newform of circuit integration replacing LSIs, the new form of circuitintegration can of course be used for the integration of the frequencyconverter 100, the sampler 101, the quadrature demodulator 102, thelow-pass filter 103, the sampling signal generator 104 and the receiveddata reproducing section 105.

Similarly, such a new form of circuit integration can be used for theintegration of the frequency converter 200, the sampler 201, thequadrature demodulator 202, the low-pass filter 203, the sampling signalgenerator 204 and the received data reproducing section 205.

Such a derivative technology may possibly be an application ofbiotechnology, for example.

Second Embodiment

In the first embodiment, a modulated low-frequency signal obtained byconverting the frequency of a modulated high-frequency signal issampled, and then an in-phase component sampled signal and a quadraturecomponent sampled signal whose phases are different from each other byπ/2 are outputted by the quadrature demodulator, which are then low-passfiltered using a low-pass filter, thereby obtaining the received data. Asecond embodiment of the present invention is directed to a wirelessdigital receiver using a complex filter instead of the quadraturedemodulator and the low-pass filter to obtain received data. The overallsystem configuration of the second embodiment is similar to that of thefirst embodiment, and thus FIG. 1 will be relied upon also in the secondembodiment.

FIG. 13 is a block diagram showing a functional configuration of thefirst wireless digital receiver 21 according to the second embodiment ofthe present invention. In FIG. 13, the first wireless digital receiver21 includes a frequency converter 600, a sampler 601, the complex filter602, a sampling signal generator 603 and a received data reproducingsection 604. The complex filter 602 and the received data reproducingsection 604 will be hereinafter referred to collectively as a“demodulation digital circuit”.

In the first wireless digital receiver 21, the frequency converter 600converts the frequency of the modulated high-frequency signal R(t) intoa modulated low-frequency signal L(t) whose center frequency is fi. Thesampler 601 samples the modulated low-frequency signal L(t) with asampling signal whose sampling frequency is fs outputted from thesampling signal generator 603 to output a sampled signal S(mTs) . Theoperation hitherto is similar to that of the first embodiment.

Therefore, the spectrum of the sampled signal S(mTs) outputted from thesampler 601 is the same as that in a case where the center frequency ofthe modulated low-frequency signal L(t) is fi=3.072 [MHz] and thesampling frequency is fs=24.576 [MHz], i.e., that shown in FIG. 8.Therefore, FIG. 8 will be relied upon also in the second embodiment.

In FIG. 8, spectra equivalent to the spectrum 300, being the spectrum ofthe modulated low-frequency signal L(t), are those spaced apart from thespectrum 300 by an integer multiple of the sampling frequency fs=24.576[MHz]. Thus, the spectrum 300 and the spectrum whose center frequency is−3.072 [MHz] are spectra having different characteristics. In order toobtain received data, the spectrum 300 or a spectrum having a spectrumspaced apart from the spectrum 300 by an integer multiple of thesampling frequency fs as a frequency component should be extracted. FIG.14 is a diagram showing exemplary pass-band characteristics of thecomplex filter 602. Where the complex filter 602 having pass-bandcharacteristics as shown in FIG. 14 is used, Ib(mTs) being an in-phasecomponent of the quadrature demodulated signal outputted from thecomplex filter 602 and Qb(mTs) being a quadrature component thereof aresignal shaving the spectrum 300 as a frequency component and whosephases are different from each other by π/2 [rad]. Although thequadrature demodulated signals Ib(mTs) and Qb(mTs) are signals whosecenter frequencies are not zero, the received data reproducing section604 can output the received data by means of delay detection, or thelike.

In the second embodiment, the configuration of the second wirelessdigital receiver 31 in the mobile station is similar to theconfiguration of the first wireless digital receiver 21, and thus FIG.13 will be relied upon also for the configuration of the second wirelessdigital receiver 31.

The second wireless digital receiver 31 is different from the firstwireless digital receiver 21 in that the modulated high-frequency signalR(t) is converted by the frequency converter 600 to the modulatedlow-frequency signal L(t) whose center frequency is fd and in that afilter that extracts the spectrum 502 whose center frequency is −9.152[MHz] as shown in FIG. 10 is used as the complex filter. Otherwise, thefirst wireless digital receiver 21 is the same as the second wirelessdigital receiver 31.

Thus, in the second embodiment, the received data can be obtained onlyby changing the pass-band characteristics of the complex filter, thusobtaining effects similar to those of the first embodiment.

The above description is directed to a case where fs=3.072 [MHz] andfs=24.576 [MHz]. Also where fi=3.072 [MHz] and fs=12.288 [MHz], similarresults can be obtained only by changing the center frequency of thepass-band characteristics of the complex filter 602 from 3.072 [MHz] to3.136 [MHz] (see FIG. 12). Also where fi=3.584 [MHz] and fs=28.672[MHz], similar results can be obtained only by changing the centerfrequency of the pass-band characteristics of the complex filter 602from 3.072 [MHz] to 3.584 [MHz] (see FIG. 15A). Also where fi=4.096[MHz] and fs=32.768 [MHz], similar results can be obtained only bychanging the center frequency of the pass-band characteristics of thecomplex filter 602 from 3.072 [MHz] to 4.096 [MHz] (see FIG. 15B). Alsowhere fi=4.608 [MHz] and fs=36.864 [MHz], similar results canbe obtainedonly by changing the center frequency of the pass-band characteristicsof the complex filter 602 from 3.072 [MHz] to 4.608 [MHz] (see FIG.15C).

Also where the center frequency of the modulated low-frequency signalL(t) inputted to the sampler 601 is fd=40.000 [MHz] and fs=24.576 [MHz]or fs=12.288 [MHz], similar results can be obtained only by changing thecenter frequency of the pass-band characteristics of the complex filter602 to −9.152 [MHz] or 3.136 [MHz], respectively. Also where fd=40.000[MHz] and fs=28.672 [MHz], similar results can be obtained only bychanging the center frequency of the pass-band characteristics of thecomplex filter 602 to 11.328 [MHz] (see FIG. 15D). Also where fd=40.000[MHz] and fs=32.768 [MHz], similar results can be obtained only bychanging the center frequency of the pass-band characteristics of thecomplex filter 602 to 7.232 [MHz] (see FIG. 15E) Also where fd=40.000[MHz] and fs=36.864 [MHz], similar results can be obtained only bychanging the center frequency of the pass-band characteristics of thecomplex filter 602 to 3.136 [MHz] (see FIG. 15F).

As for the complex filter characteristics, in a case where an FIR(Finite Impulse Response) filter is used as the complex filter, forexample, the number of taps can be determined in advance so as toaccommodate any of the center frequencies of the pass-bandcharacteristics of 3.072 [MHz], 3.136 [MHz] and −9.152 [MHz], wherebyall of the cases mentioned above can be addressed only by selecting anappropriate tap coefficient. Thus, by using an FIR with which one ofdifferent tap coefficients can be selected, the same demodulationdigital circuit, being a complex filter, can be used for the mobilestation and for the base station, whereby it is possible to reduce thecost.

Note that in FIG. 13 being a block diagram showing a functionalconfiguration of the first wireless digital receiver 21 according to thesecond embodiment of the present invention, the frequency converter 600,the sampler 601, the complex filter 602, the sampling signal generator603 and the received data reproducing section 604 are typically eachimplemented in the form of an LSI being an integrated circuit. Thesecomponents may be individually formed into a separate chip, or some orall of them may be formed together into a single chip.

While the term “LSI” is used herein as the type of integrated circuitused in the present invention, integrated circuits are also called“ICs”, “system LSIs”, “super LSIs” or “ultra LSIs” depending on thedegree of integration. Moreover, the form of an integrated circuit thatcan be used with the present invention is not limited to an LSI, but mayalternatively be a dedicated circuit or a general-purpose processor. Itmay alternatively be an FPGA (Field Programmable Gate Array) beingprogrammable after the LSI is manufactured, or a reconfigurableprocessor in which the interconnections and settings of circuit cells inthe LSI can be reconfigured. Furthermore, if advancements in thesemiconductor technology or derivative technologies bring forth a newform of circuit integration replacing LSIs, the new form of circuitintegration can of course be used for the integration of the frequencyconverter 600, the sampler 601, the complex filter 602, the samplingsignal generator 603 and the received data reproducing section 604. Sucha derivative technology may possibly be an application of biotechnology,for example.

Third Embodiment

FIG. 1 will be relied upon also in a third embodiment of the presentinvention. FIG. 16A and FIG. 16B are block diagrams each showing afunctional configuration of the first wireless digital receiver 21according to the third embodiment of the present invention.

In FIG. 16A, the first wireless digital receiver 21 includes a frequencyconverter 800, a sampler 801, the quadrature demodulator 802, anautomatic frequency controller 803, a low-pass filter 804, a samplingsignal generator 805, a detector 806 and a data determination section807. The quadrature demodulator 802, the automatic frequency controller803, the low-pass filter 804, the detector 806 and the datadetermination section 807 will be hereinafter referred to collectivelyas a “demodulation digital circuit”.

In the third embodiment, the frequency converter 800 does not convertthe modulated high-frequency signal R(t) to a modulated low-frequencysignal whose center frequency is 3.072 [MHz]. The following descriptionis directed to a case where the frequency converter 800 converts themodulated high-frequency signal R(t) to the modulated low-frequencysignal L(t) whose center frequency is fj=3.000 [MHz].

The sampler 801 samples the modulated low-frequency signal L(t) insynchronism with the sampling signal whose frequency is fs=24.576 [MHz]outputted from the sampling signal generator 805 to output the sampledsignal S(mTs).

The quadrature demodulator 802 assumes that the center frequency of themodulated low-frequency signal L(t) is fi=3.072 [MHz], and performs anoperation S(mTs)×exp (−jθ×mTs) using θ [rad] expressed as shown inExpression 23 to obtain the in-phase component sampled signal I(mTs) andthe quadrature component sampled signal Q(mTs).

FIG. 17 is a diagram showing the spectrum of the in-phase componentsampled signal I(mTs) and the quadrature component sampled signal Q(mTs)outputted from the quadrature demodulator 802. In FIG. 17, a spectrum900 is the spectrum of the modulated low-frequency signal L(t). Theother spectra are folding spectra occurring as a result of sampling themodulated low-frequency signal L(t) with a sampling period of Ts. Acomparison between FIG. 17 and FIG. 8 shows that these spectra as awhole are shifted from each other by 0.072 [MHz], which is thedifference between 3.072 [MHz] being the intended frequency of themodulated low-frequency signal L(t) and the actual frequency 3.000 [MHz]thereof.

The automatic frequency controller 803 converts the frequency of thespectrum 900 so that it is frequency-shifted to its intended centerfrequency of 3.072 [MHz]. In other words, the automatic frequencycontroller 803 converts the entire spectrum shown in FIG. 17 so that thecenter frequency of the spectrum 900 is 3.072 [MHz]. Such an automaticfrequency controller 803 is disclosed in Japanese Patent No. 3327152,Japanese Laid-Open Patent Publication No. 6-120997, etc.

If the automatic frequency controller 803 performing such an operationis provided between the quadrature demodulator 802 and the low-passfilter 804, the low-pass filter 804 can be a filter having the samepass-band characteristics as those shown in FIG. 3. The detector 806provided on the output side of the low-pass filter 804 performs a delaydetection operation to output detection signals DETI(mTs) and DETQ(mTs)to the data determination section 807. The data determination section807 detects a phase using the signals DETI(mTs) and DETQ(mTs), andoutputs the received data based on the detected phase.

Thus, the third embodiment provides the following advantage. In a casewhere fi as calculated in the first embodiment cannot be used, e.g.,where it is necessary to order a tailored frequency oscillator in orderto use fi as calculated in the first embodiment, it is possible toobtain a sampled signal having a component whose center frequency is fiby digitally correcting the frequency with an automatic frequencycontroller by using a frequency converter capable of converting afrequency to another frequency near fi. Thus, the received data can beproperly reproduced. By providing a frequency converter using ageneral-purpose local oscillator so that a frequency can be converted toanother frequency near fi, it is possible to reduce the cost of thewireless digital receiver.

While the above description is directed to a case where the automaticfrequency controller 803 is provided immediately after the quadraturedemodulator 802, similar effects can be obtained also with aconfiguration as shown in FIG. 16B. Note however that where aconfiguration as shown in FIG. 16B is used, it is necessary to use anautomatic frequency controller as disclosed in Japanese Patent No.3088893, Japanese Laid-Open Patent Publication No. 10-98500, etc.

While the above description is directed to a case where the centerfrequency of the modulated low-frequency signal L(t) is shifted from fi,similar effects can be obtained also where the center frequency of themodulated low-frequency signal L(t) is shifted from fd. Specifically, byperforming the frequency shifting operation at the automatic frequencycontroller 803 so that the center frequency of the spectrum of themodulated low-frequency signal L(t) is equal to fd, pass-bandcharacteristics as shown in FIG. 3 can be used as the pass-bandcharacteristics of the low-pass filter 804, and the received data can beobtained by a delay detection circuit, or the like, provided on theoutput side of the low-pass filter 804.

While fj=3.000 [MHz] in the above description, the present invention isnot limited to this as long as the frequency shift Δf between fi=3.072and fj satisfies |Δf|<0.512 [MHz]. The reason for this will now bedescribed. A frequency is an amount of phase change per unit time.Therefore, there is a one-to-one correspondence between a frequencyshift and a phase shift. The DSRC system uses a format of transmitteddata in which the beginning portion of each frame contains a preamblepattern made up of symbols each having a phase different from that ofthe next symbol by π. By using the preamble pattern, a phase correctionof up to ±π/2 (excluding ±π/2) can be performed in principle. Where aphase difference of π/2 is converted to a frequency, the symbol datarate fsym is involved in the conversion formula, whereby the frequencyfor π/2 varies depending on the value of fsym. This is expressed inExpression 30. $\begin{matrix}{{\theta\quad{err}} = {2\pi \times {\frac{\Delta\quad f}{fs}\lbrack{rad}\rbrack}}} & {{Exp}.\quad 30}\end{matrix}$Herein, θerr is the phase for the frequency shift Δf. In the presentembodiment, fsym=2.048 [MHz]. Therefore, where θerr=π/2 and fs=2.048[MHz], Expression 30 can be rearranged with respect to Δf to yield|Δf|<0.512 [MHz].

The above description is directed to a case where the automaticfrequency controller 803 being a circuit for correcting a signal whosecenter frequency is fj to a signal whose center frequency is fi isprovided following the sampler 801. Alternatively, such a frequencycorrection circuit for correcting a frequency may be provided precedingthe sampler 801. Thus, the low-frequency signal downconverted by thefrequency converter 800 may be demodulated after being corrected to asignal whose center frequency is fi at a position either preceding orfollowing the sampler 801.

Note that in FIG. 16A being a block diagram showing a functionalconfiguration of the first wireless digital receiver 21 according to thethird embodiment of the present invention, the frequency converter 800,the sampler 801, the quadrature demodulator 802, the automatic frequencycontroller 803, the low-pass filter 804, the sampling signal generator805, the detector 806 and the data determination section 807 aretypically each implemented in the form of an LSI being an integratedcircuit. These components may be individually formed into a separatechip, or some or all of them may be formed together into a single chip.

Also where a configuration as shown in FIG. 16B is used, the frequencyconverter 800, the sampler 801, the quadrature demodulator 802, theautomatic frequency controller 803, the low-pass filter 804, thesampling signal generator 805, the detector 806 and the datadetermination section 807 may be individually formed into a separatechip, or some or all of them may be formed together into a single chip.

While the term “LSI” is used herein as the type of integrated circuitused in the present invention, integrated circuits are also called“ICs”, “system LSIs”, “super LSIs” or “ultra LSIs” depending on thedegree of integration. Moreover, the form of an integrated circuit thatcan be used with the present invention is not limited to an LSI, but mayalternatively be a dedicated circuit or a general-purpose processor. Itmay alternatively be an FPGA (Field Programmable Gate Array) beingprogrammable after the LSI is manufactured, or a reconfigurableprocessor in which the interconnections and settings of circuit cells inthe LSI can be reconfigured. Furthermore, if advancements in thesemiconductor technology or derivative technologies bring forth a newform of circuit integration replacing LSIs, the new form of circuitintegration can of course be used for the integration of the frequencyconverter 800, the sampler 801, the quadrature demodulator 802, theautomatic frequency controller 803, the low-pass filter 804, thesampling signal generator 805, the detector 806 and the datadetermination section 807.

Also where a configuration as shown in FIG. 16B is used, such a new formof circuit integration replacing LSIs brought forth by advancements inthe semiconductor technology or derivative technologies may be used forthe integration of the frequency converter 800, the sampler 801, thequadrature demodulator 802, the automatic frequency controller 803, thelow-pass filter 804, the sampling signal generator 805, the detector 806and the data determination section 807.

Such a derivative technology may possibly be an application ofbiotechnology, for example.

Fourth Embodiment

A fourth embodiment of the present invention is directed to abase-station wireless communications device obtained by combiningtogether the first wireless transmitter and the first wireless digitalreceiver in the base station, and a mobile-station wirelesscommunications device obtained by combining together the second wirelesstransmitter and the second wireless digital receiver in the mobilestation.

FIG. 18 is a diagram showing a configuration of a base-station wirelesscommunications device 12 according to the fourth embodiment of thepresent invention. In FIG. 18, the base-station wireless communicationsdevice 12 includes an antenna 1200, a band-pass filter 1216, atransmission/reception selector switch 1211, an amplifier 1201, a firstmixer 1202, a second mixer 1203, a first local oscillator 1206, a firstlow-pass filter 1204, a second low-pass filter 1205, a first sampler1207, a second sampler 1208, a sampling signal generator 1209, ademodulation digital circuit 1210, a transmission high-frequency circuit1212, a third mixer 1213, a second local oscillator 1214 and atransmitter circuit 1215.

In the base-station wireless communications device 12, thesignal-receiving operation is performed by using the antenna 1200, theband-pass filter 1216, the transmission/reception selector switch 1211,the amplifier 1201, the first mixer 1202, the second mixer 1203, thefirst local oscillator 1206, the first low-pass filter 1204, the secondlow-pass filter 1205, the first sampler 1207, the second sampler 1208,the sampling signal generator 1209 and the demodulation digital circuit1210. The signal-transmitting operation is performed by using thetransmitter circuit 1215, the second local oscillator 1214, the thirdmixer 1213, the transmission high-frequency circuit 1212, thetransmission/reception selector switch 1211, the band-pass filter 1216and the antenna 1200.

In the signal-receiving operation, the transmission/reception selectorswitch 1211 is switched so that the antenna 1200 and the amplifier 1201are connected to eachother. The modulated high-frequency signal R(t)received by the antenna 1200 from the mobile station whose centerfrequency is fc is first passed through the band-pass filter 1216 toremove signals of frequency bands that are used neither in the basestation nor in the mobile station, and is then inputted to the amplifier1201. The amplifier 1201 amplifies the modulated high-frequency signalR(t) to an appropriate level, and inputs the amplified signal to thefirst mixer 1202 and the second mixer 1203. The first local oscillator1206 outputs a sine wave whose center frequency is fc-fi. Herein, fi is3.072 [MHz] as calculated in the first embodiment.

The first mixer 1202 multiplies the sine wave outputted from the firstlocal oscillator 1206 whose center frequency is fc-fi with the modulatedhigh-frequency signal R(t) to output a modulatedlow-to-intermediate-frequency signal in-phase component RXI(t) whosecenter frequency is fi. The first low-pass filter 1204 removes ahigh-frequency component from the modulatedlow-to-intermediate-frequency signal in-phase component RXI(t), andpasses the filtered signal to the first sampler 1207.

The second mixer 1203 multiplies a signal outputted from the first localoscillator 1206 whose center frequency is fc-fi and whose phase isshifted from that of the sine wave by π/2 with the modulatedhigh-frequency signal R(t) to output a modulatedlow-to-intermediate-frequency signal quadrature component RXQ(t) whosecenter frequency is fi. The second low-pass filter 1205 removes ahigh-frequency component from the modulated low-to intermediatefrequency signal quadrature component RXQ(t), and passes the filteredsignal to the second sampler 1208.

The first sampler 1207 samples the modulatedlow-to-intermediate-frequency signal in-phase component RXI(t) insynchronism with a signal outputted from the sampling signal generator1209 whose frequency is fs=24.576 [MHz] to output the in-phase componentsampled signal I(mTs).

The second sampler 1208 samples the modulatedlow-to-intermediate-frequency signal quadrature component RXQ(t) insynchronism with a signal outputted from the sampling signal generator1209 whose frequency is fs=24.576 [MHz] to output the quadraturecomponent sampled signal Q(mTs).

The demodulation digital circuit 1210 receives the in-phase componentsampled signal I(mTs) and the quadrature component sampled signalQ(mTs), and performs a quadrature demodulation operation on the receivedsignals. Then, the demodulation digital circuit 1210 low-pass-filtersthe demodulated signals to output received data.

In FIG. 18, the first and second mixers 1202 and 1203, the first localoscillator 1206 and the first and second low-pass filters 1204 and 1205correspond to the frequency converter 100 in the first embodiment. Thefirst and second samplers 1207 and 1208 correspond to the sampler 101illustrated in the first embodiment. In the fourth embodiment,quadrature data is sampled, unlike in the first embodiment. However, thefourth embodiment is substantially the same as the first embodimentsince the values of fi and fs used in the first embodiment are used alsoin the fourth embodiment. The sampling signal generator 1209 correspondsto the sampling signal generator 104 illustrated in the firstembodiment. The demodulation digital circuit 1210 corresponds to thequadrature demodulator 102, the low-pass filter 103 and the receiveddata reproducing section 105 in the first embodiment.

In the signal-transmitting operation, data to be transmitted ismodulated according to the π/4 shift QPSK scheme in the transmittercircuit 1215, and is outputted as a transmitted signal B(t). The thirdmixer 1213 multiplies the transmitted signal B(t) by a signal outputtedfrom the local oscillator 1214 whose center frequency is fc-fd to outputa modulated high-frequency signal TX(t). The modulated high-frequencysignal TX(t) is passed through the transmission high-frequency circuit1212 to remove unnecessary frequency components, and adjusted to anappropriate transmission power level, after which the signal is radiatedoff the antenna 1200 in the form of a radio wave.

FIG. 19 is a diagram showing a configuration of a mobile-stationwireless communications device 11 according to the fourth embodiment ofthe present invention. In FIG. 19, the mobile-station wirelesscommunications device 11 includes an antenna 1100, a band-pass filter1112, a transmission/reception selector switch 1108, an amplifier 1101,a first mixer 1102, a local oscillator 1103, a low-pass filter 104, asampler 1105, a sampling signal generator 1106, a demodulation digitalcircuit 1107, a transmission high-frequency circuit 1109, a second mixer1110 and a transmitter circuit 1111.

In the mobile-station wireless communications device, thesignal-receiving operation is performed by using the antenna 1100, theband-pass filter 1112, the transmission/reception selector switch 1108,the amplifier 1101, the first mixer 1102, the local oscillator 1103, thelow-pass filter 1104, the sampler 1105, the sampling signal generator1106 and the demodulation digital circuit 1107. The signal-transmittingoperation is performed by using the transmitter circuit 1111, the secondmixer 1110, the local oscillator 1103, the transmission high-frequencycircuit 1109, the transmission/reception selector switch 1108, theband-pass filter 1112 and the antenna 1100.

In the signal-receiving operation, the transmission/reception selectorswitch 1108 is switched so that the antenna 1100 and the amplifier 1101are connected to each other. The modulated high-frequency signal RL(t)from the base station received by the antenna 1100 whose centerfrequency is fc-fd is first passed through the band-pass filter 1112 toremove signals of frequency bands that are used neither in the basestation nor in the mobile station, and is then inputted to the amplifier1101. The amplifier 1101 amplifies the modulated high-frequency signalRL(t) to an appropriate level, and inputs the amplified signal to thefirst mixer 1102. The first local oscillator 1103 outputs a sine wavewhose center frequency is fc.

The first mixer 1102 multiplies the sine wave outputted from the localoscillator 1103 whose center frequency is fc with the modulatedhigh-frequency signal RL(t) to output a modulatedlow-to-intermediate-frequency signal L(t) whose center frequency is fdto the low-pass filter 1104. In the DSRC system, the frequencydifference between the downlink and the uplink is 40.000 [MHz]Therefore, fd=40.000 [MHz]. The low-pass filter 1104 removes ahigh-frequency component from the modulatedlow-to-intermediate-frequency signal L(t), and passes the filteredsignal to the sampler 1105.

The sampler 1105 samples the modulated low-to-intermediate-frequencysignal L(t) in synchronism with a signal outputted from the samplingsignal generator 1106 whose frequency is fs=24.576 [MHz] to output asampled signal Ls(mTs).

The demodulation digital circuit 1107 receives the sampled signalLs(mTs), and performs a quadrature demodulation operation on thereceived signal. Then, the demodulation digital circuit 1107low-pass-filters the demodulated signal to output received data.

In FIG. 19, the first mixer 1102, the local oscillator 1103 and thelow-pass filter 1104 correspond to the frequency converter 200 in thefirst embodiment. The sampler 1105 corresponds to the sampler 201 in thefirst embodiment. The sampling signal generator 1106 corresponds to thesampling signal generator 204 in the first embodiment. The demodulationdigital circuit 1107 corresponds to the quadrature demodulator 202, thelow-pass filter 203 and the received data reproducing section 205 in thefirst embodiment.

In the signal-transmitting operation, data to be transmitted ismodulated according to the π/4 shift QPSK scheme in the transmittercircuit 1111, and is outputted as the transmitted signal B(t). Thesecond mixer 1110 multiplies the transmitted signal B(t) by a signaloutputted from the local oscillator 1103 whose center frequency is fc tooutput a modulated high-frequency signal TX(t). The modulatedhigh-frequency signal TX(t) is passed through the transmissionhigh-frequency circuit 1109 to remove unnecessary frequency components,and adjusted to an appropriate transmission power level, after which thesignal is radiated off the antenna 1100 in the form of a radio wave.

Thus, in the fourth embodiment, the same sampling frequency is used forthe mobile station and for the base station, and thus the samedemodulation digital circuit can be used for the mobile station and forthe base station, whereby it is possible to provide a wirelesscommunications system and a wireless digital receiver for use therein ata low cost.

Note that the components of the base-station wireless communicationsdevice 12 shown in FIG. 18, i.e., the antenna 1200, the band-pass filter1216, the transmission/reception selector switch 1211, the amplifier1201, the first mixer 1202, the second mixed 1203, the first localoscillator 1206, the first low-pass filter 1204, the second low-passfilter 1205, the first sampler 1207, the second sampler 1208, thesampling signal generator 1209, the demodulation digital circuit 1210,the transmission high-frequency circuit 1212, the third mixer 1213, thesecond local oscillator 1214 and the transmitter circuit 1215, aretypically each implemented in the form of an LSI being an integratedcircuit. These components may be individually formed into a separatechip, or some or all of them may be formed together into a single chip.

Similarly, the components of the mobile-station wireless communicationsdevice 11 shown in FIG. 19, i.e., the antenna 1100, the band-pass filter1112, the transmission/reception selector switch 1108, the amplifier1101, the first mixer 1102, the local oscillator 1103, the low-passfilter 1104, the sampler 1105, the sampling signal generator 1106, thedemodulation digital circuit 1107, the transmission high-frequencycircuit 1109, the second mixer 1110 and the transmitter circuit 1111,are typically each implemented in the form of an LSI being an integratedcircuit. These components may be individually formed into a separatechip, or some or all of them may be formed together into a single chip.

While the term “LSI” is used herein as the type of integrated circuitused in the present invention, integrated circuits are also called“ICs”, “system LSIs”, “super LSIs” or “ultra LSIs” depending on thedegree of integration. Moreover, the form of an integrated circuit thatcan be used with the present invention is not limited to an LSI, but mayalternatively be a dedicated circuit or a general-purpose processor. Itmay alternatively be an FPGA (Field Programmable Gate Array) beingprogrammable after the LSI is manufactured, or a reconfigurableprocessor in which the interconnections and settings of circuit cells inthe LSI can be reconfigured. Furthermore, if advancements in thesemiconductor technology or derivative technologies bring forth a newform of circuit integration replacing LSIs, the new form of circuitintegration can of course be used for the integration of the antenna1200, the band-pass filter 1216, the transmission/reception selectorswitch 1211, the amplifier 1201, the first mixer 1202, the second mixer1203, the first local oscillator 1206, the first low-pass filter 1204,the second low-pass filter 1205, the first sampler 1207, the secondsampler 1208, the sampling signal generator 1209, the demodulationdigital circuit 1210, the transmission high-frequency circuit 1212, thethird mixer 1213, the second local oscillator 1214 and the transmittercircuit 1215.

Similarly, such a new form of circuit integration replacing LSIs broughtforth by advancements in the semiconductor technology or derivativetechnologies may be used for the integration of the antenna 1100, theband-pass filter 1112, the transmission/reception selector switch 1108,the amplifier 1101, the first mixer 1102, the local oscillator 1103, thelow-pass filter 1104, the sampler 1105, the sampling signal generator1106, the demodulation digital circuit 1107, the transmissionhigh-frequency circuit 1109, the second mixer 1110 and the transmittercircuit 1111.

Such a derivative technology may possibly be an application ofbiotechnology, for example.

While the DSRC system has been described in detail in the first tofourth embodiments, it is understood that a wireless communicationssystem and a wireless data receiver providing similar effects can beobtained with other types of FDD systems.

The various functional blocks mentioned above in the first to fourthembodiments may be any means capable of performing their functions. Forexample, the frequency converter may be any frequency converting means,the sampler may be any sampling means, the demodulation digital circuitmay be any digital demodulating means, the quadrature demodulator may beany quadrature demodulating means, the low-pass filter may be anylow-pass filtering means, the received data reproducing section may beany received data reproducing means, and the complex filter may be anycomplex filtering means. These functional blocks are not limited to anyparticular types of devices as long as they are operable to performtheir functions.

While the invention has been described in detail, the foregoingdescription is in all aspects illustrative and not restrictive. It isunderstood that numerous other modifications and variations can bedevised without departing from the scope of the invention.

INDUSTRIAL APPLICABILITY

A wireless communications system and a wireless digital receiver for usetherein according to the present invention can be provided at a lowcost, and are useful in various applications such as a wirelesscommunications application using an FDD architecture.

1. A wireless communications system for transmitting/receiving a firstwireless signal from a first wireless communications device and a secondwireless signal from a second wireless communications device, the firstand second wireless signals having different frequency bands from eachother, wherein: the first wireless communications device includes: afirst frequency converter operable to downconvert the second wirelesssignal transmitted from the second wireless communications device to afirst low-frequency signal; a first sampler for operable to oversamplethe first low-frequency signal downconverted by the first frequencyconverter; and a first demodulation digital circuit operable todemodulate the signal oversampled by the first sampler; the signaldemodulated by the first demodulation digital circuit has a centerfrequency of fi [Hz]; the second wireless communications deviceincludes: a second frequency converter operable to downconvert the firstwireless signal transmitted from the first wireless communicationsdevice to a second low-frequency signal whose center frequency fd [Hz]is equal to a difference between a center frequency of the firstwireless signal and that of the second wireless signal; a second sampleroperable to undersample the second low-frequency signal downconverted bythe second frequency converter; and a second demodulation digitalcircuit operable to demodulate the signal undersampled by the secondsampler; a sampling frequency used in the first sampler and that used inthe second sampler are the same sampling frequency fs [Hz]; the samplingfrequency fs [Hz] is set to a value that is an even-number multiple of awireless symbol transmission rate such that oversampling is done in thefirst sampler and undersampling is done in the second sampler; and thecenter frequency fi [Hz] is ½ to 1 times a frequency corresponding to abandwidth of the first and second wireless signals and is ½^(N) (N is anatural number) times the sampling frequency fs [Hz].
 2. The wirelesscommunications system according to claim 1, wherein where the bandwidthof the first and second wireless signals is 2×Bch [Hz] and the wirelesssymbol transmission rate is fsym [Hz], the sampling frequency fs [Hz]and the center frequency fi [Hz] are expressed as shown in the followingexpressions: $\begin{matrix}{{fi} = \frac{2k\quad{fsym}}{2^{N}}} \\{{fs} = {2^{N}{fi}}}\end{matrix}$ where k is an integer satisfying $\begin{matrix}{{\frac{{fd} + {Bch}}{\left( {n + 1} \right){fsym}} \leqq k \leqq \frac{{fd} - {Bch}}{n\quad{fsym}}}{and}} & {{Exp}.\quad 12} \\{k \leqq \frac{fd}{2{fsym}}} & {{Exp}.\quad 14}\end{matrix}$ and N is an integer satisfying $\begin{matrix}{{\log\quad 2\left\{ \frac{{fd} + {Bch}}{\left( {n + 1} \right){Bch}} \right\}} \leqq N \leqq {\log\quad 2\left\{ \frac{2\left( {{fd} - {Bch}} \right)}{n\quad{Bch}} \right\}}} & {{Exp}.\quad 22}\end{matrix}$ where n is an integer satisfying
 3. The wirelesscommunications system according to claim 1, wherein: $\begin{matrix}{1 \leqq n \leqq \frac{{fd} - {Bch}}{2{Bch}}} & {{Exp}.\quad 7}\end{matrix}$ the first frequency converter downconverts the secondwireless signal transmitted from the second wireless communicationsdevice to a first low-frequency signal whose center frequency is fj[Hz]; and the first low-frequency signal is demodulated by the firstdemodulation digital circuit after being corrected to a signal whosecenter frequency is fi [Hz] at a position preceding or following thefirst sampler.
 4. The wireless communications system according to claim1, wherein: the center frequency fd is 40.000 [MHz]; and the frequencyfi and the sampling frequency fs are fi=3.072 [MHz] and fs=24.576 [MHz],fi=3.072 [MHz] and fs=12.288 [MHz], fi=4.608 [MHz] and fs=36.864 [MHz],fi=4.096 [MHz] and fs=32.768 [MHz], or fi=3.584 [MHz] and fs=28.672[MHz].
 5. The wireless communications system according to claim 1,wherein: the first demodulation digital circuit includes: a firstquadrature demodulator operable to quadrature-demodulate the signaloversampled by the first sampler; a first low-pass filter for low passfiltering operable to low-pass-filter the signal quadrature-demodulatedby the first quadrature demodulator; and a first received datareproducing section operable to reproduce received data from the signallow-pass-filtered by the first low-pass filter; the second demodulationdigital circuit includes: a second quadrature demodulator operable toquadrature-demodulate the signal undersampled by the second sampler; asecond low-pass filter operable to low-pass-filter the signalquadrature-demodulated by the second quadrature demodulator; and asecond received data reproducing section operable to reproduce receiveddata from the signal low-pass-filtered by the second low-pass filter;the first quadrature demodulator converts the signal oversampled by thefirst sampler to a signal including a component whose center frequencyis zero; and the second quadrature demodulator converts the signalundersampled by the second sampler to a signal including a componentwhose center frequency is zero.
 6. The wireless communications systemaccording to claim 1, wherein: the first demodulation digital circuitincludes: a first complex filter operable to filter, by using a digitalfilter, either one of a positive frequency component and a negativefrequency component of the signal oversampled by the first sampler whosecenter frequency is closer to zero; and a first received datareproducing section operable to reproduce received data from the signalfiltered by the first complex filter; and the second demodulationdigital circuit includes: a second complex filter operable to filter, byusing a digital filter, either one of a positive frequency component anda negative frequency component of the signal undersampled by the secondsampler whose center frequency is closer to zero; and a second receiveddata reproducing section operable to reproduce received data from thesignal filtered by the second complex filter.
 7. The wirelesscommunications system according to claim 3, wherein: the firstdemodulation digital circuit includes: a first quadrature demodulatoroperable to quadrature-demodulate the signal oversampled by the firstsampler; a first low-pass filter operable to low-pass-filter the signaloutputted from the first quadrature demodulator; and a first receiveddata reproducing section operable to reproduce received data from thesignal low-pass-filtered by the first low-pass filter; the seconddemodulation digital circuit includes: a second quadrature demodulatoroperable to quadrature-demodulate the signal undersampled by the secondsampler; a second low-pass filter operable to low-pass-filter the signalquadrature-demodulated by the second quadrature demodulator; and asecond received data reproducing section operable to reproduce receiveddata from the signal low-pass-filtered by the second low-pass filter;the first quadrature demodulator converts the signal oversampled by thefirst sampler to a signal including a component whose center frequencyis zero; and the second quadrature demodulator converts the signalundersampled by the second sampler to a signal including a componentwhose center frequency is zero.
 8. The wireless communications systemaccording to claim 7, wherein the frequency fj [Hz] is 3.000 [MHz].
 9. Awireless digital receiver in a wireless communications system fortransmitting/receiving a first wireless signal from a first wirelesscommunications device and a second wireless signal from a secondwireless communications device, the first and second wireless signalshaving different frequency bands from each other, the wireless digitalreceiver receiving the second wireless signal in the first wirelesscommunications device and digitally demodulating the second wirelesssignal, the wireless digital receiver comprising: a frequency converteroperable to downconvert the second wireless signal transmitted from thesecond wireless communications device to a low-frequency signal whosecenter frequency is fi [Hz]; a sampler operable to oversample thelow-frequency signal downconverted by the frequency converter; and ademodulation digital circuit operable to demodulate the signaloversampled by the sampler, wherein: a sampling frequency used in thesampler and that used in the second wireless communications device arethe same sampling frequency fs [Hz]; the sampling frequency fs [Hz] isset to a value that is an even-number multiple of a wireless symboltransmission rate such that oversampling is done in the sampler andundersampling is done in a sampler of the second wireless communicationsdevice; and the center frequency fi [Hz] of the low-frequency signal is½ to 1 times a frequency corresponding to a bandwidth of the first andsecond wireless signals and is ½^(N) (N is a natural number) times thesampling frequency fs [Hz].
 10. The wireless digital receiver accordingto claim 9, wherein where the bandwidth of the first and second wirelesssignals is 2×Bch [Hz] and the wireless symbol transmission rate is fsym[Hz], the sampling frequency fs [Hz] and the center frequency fi [Hz] ofthe low-frequency signal are expressed as shown in the followingexpressions: $\begin{matrix}{{fi} = \frac{2{kfsym}}{2^{N}}} \\{{fs} = {2^{N}{fi}}}\end{matrix}$ where k is an integer satisfying $\begin{matrix}{{\frac{{fd} + {Bch}}{\left( {n + 1} \right){fsym}} \leqq k \leqq \frac{{fd} - {Bch}}{n\quad{fsym}}}{and}} & {{Exp}.\quad 12} \\{k \leqq \frac{fd}{2{fsym}}} & {{Exp}.\quad 14}\end{matrix}$ and N is an integer satisfying $\begin{matrix}{{\log\quad 2\left\{ \frac{{fd} + {Bch}}{\left( {n + 1} \right){Bch}} \right\}} \leqq N \leqq {\log\quad 2\left\{ \frac{2\left( {{fd} - {Bch}} \right)}{n\quad{Bch}} \right\}}} & {{Exp}.\quad 22}\end{matrix}$ where n is an integer satisfying $\begin{matrix}{1 \leqq n \leqq \frac{{fd} - {Bch}}{2{Bch}}} & {{Exp}.\quad 7}\end{matrix}$
 11. The wireless digital receiver according to claim 9,wherein the center frequency fi and the sampling frequency fs arefi=3.072 [MHz] and fs=24.576 [MHz], fi=3.072 [MHz] and fs=12.288 [MHz],fi=4.608 [MHz] and fs=36.864 [MHz], fi=4.096 [MHz] and fs=32.768 [MHz],or fi=3.584 [MHz] and fs=28.672 [MHz].
 12. The wireless digital receiveraccording to claim 9, wherein: the demodulation digital circuitincludes: a quadrature demodulator operable to quadrature-demodulate thesignal oversampled by the sampler; a low-pass filter operable tolow-pass-filter the signal quadrature-demodulated by the quadraturedemodulator; and a received data reproducing section operable toreproduce received data from the signal low-pass-filtered by thelow-pass filter; and the quadrature demodulator converts the signaloversampled by the sampler to a signal including a component whosecenter frequency is zero.
 13. The wireless digital receiver according toclaim 9, wherein the demodulation digital circuit includes: a complexfilter operable to filter, by using a digital filter, either one of apositive frequency component and a negative frequency component of thesignal oversampled by the sampler whose center frequency is closer tozero; and a received data reproducing section operable to reproducereceived data from the signal filtered by the complex filter.
 14. Awireless digital receiver in a wireless communications system fortransmitting/receiving a first wireless signal from a first wirelesscommunications device and a second wireless signal from a secondwireless communications device, the first and second wireless signalshaving different frequency bands from each other, the wireless digitalreceiver receiving the first wireless signal in the second wirelesscommunications device and digitally demodulating the first wirelesssignal, the wireless digital receiver comprising: a frequency converteroperable to downconvert the first wireless signal transmitted from thefirst wireless communications device to a low-frequency signal whosecenter frequency fd [Hz] is equal to a difference between a centerfrequency of the first wireless signal and that of the second wirelesssignal; a sampler operable to undersample the low-frequency signaldownconverted by the frequency converter; and a demodulation digitalcircuit operable to demodulate the signal undersampled by the sampler,wherein: a sampling frequency used in the sampler and that used in thefirst wireless communications device are the same sampling frequency fs[Hz]; and the sampling frequency fs [Hz] is set to a value that is aneven-number multiple of a wireless symbol transmission rate such thatundersampling is done in the sampler and oversampling is done in asampler of the first wireless communications device.
 15. The wirelessdigital receiver according to claim 14, wherein where the bandwidth ofthe first and second wireless signals is 2×Bch [Hz] and the wirelesssymbol transmission rate is fsym [Hz], the sampling frequency fs [Hz] isexpressed as shown in the following expression:fs=2kfsym where k is an integer satisfying $\begin{matrix}{{\frac{{fd} + {Bch}}{\left( {n + 1} \right){fsym}} \leqq k \leqq \frac{{fd} - {Bch}}{n\quad{fsym}}}{and}} & {{Exp}.\quad 12} \\{k \leqq \frac{fd}{2{fsym}}} & {{Exp}.\quad 14}\end{matrix}$ where n is an integer satisfying $\begin{matrix}{1 \leqq n \leqq \frac{{fd} - {Bch}}{2{Bch}}} & {{Exp}.\quad 7}\end{matrix}$
 16. The wireless digital receiver according to claim 14,wherein: the center frequency fd is 40.000 [MHz]; and the samplingfrequency fs is 24.576 [MHz], 12.288 [MHz], fs=36.864 [MHz], fs=32.768[MHz] or fs=28.672 [MHz].
 17. The wireless digital receiver according toclaim 14, wherein: the demodulation digital circuit includes: aquadrature demodulator operable to quadrature-demodulate the signalundersampled by the sampler; and a low-pass filter operable tolow-pass-filter the signal quadrature-demodulated by the quadraturedemodulator; and a received data reproducing section operable toreproduce received data from the signal low-pass-filtered by thelow-pass filter; and the quadrature demodulator converts the signalundersampled by the sampler to a signal including a component whosecenter frequency is zero.
 18. The wireless digital receiver according toclaim 14, wherein the demodulation digital circuit includes: a complexfilter operable to filter, by using a digital filter, either one of apositive frequency component and a negative frequency component of thesignal undersampled by the sampler whose center frequency is closer tozero; and a received data reproducing section operable to reproducereceived data from the signal filtered by the complex filter.
 19. Awireless digital receiver in a wireless communications system fortransmitting/receiving a first wireless signal from a first wirelesscommunications device and a second wireless signal from a secondwireless communications device, the first and second wireless signalshaving different frequency bands from each other, the wireless digitalreceiver receiving the second wireless signal in the first wirelesscommunications device and digitally demodulating the second wirelesssignal, the wireless digital receiver comprising: a frequency converteroperable to downconvert the second wireless signal transmitted from thesecond wireless communications device to a low-frequency signal whosecenter frequency is fj [Hz]; a sampler operable to oversample thelow-frequency signal downconverted by the frequency converter; and ademodulation digital circuit operable to demodulate the signaloversampled by the sampler after correcting a center frequency thereofto fi [Hz], wherein: a sampling frequency used in the sampler and thatused in the second wireless communications device are the same samplingfrequency fs [Hz]; the sampling frequency fs [Hz] is set to a value thatis an even-number multiple of a wireless symbol transmission rate suchthat oversampling is done in the sampler and undersampling is done in asampler of the second wireless communications device; and the centerfrequency fi [Hz] is ½ to 1 times a frequency corresponding to abandwidth of the first and second wireless signals and is ½^(N) (N is anatural number) times the sampling frequency fs [Hz].
 20. The wirelessdigital receiver according to claim 19, wherein where the bandwidth ofthe first and second wireless signals is 2×Bch [Hz] and the wirelesssymbol transmission rate is fsym [Hz], the sampling frequency fs [Hz]and the frequency fi [Hz] are expressed as shown in the followingexpressions: $\begin{matrix}{{fi} = \frac{2k\quad{fsym}}{2^{N}}} \\{{fs} = {2^{N}{fi}}}\end{matrix}$ where k is an integer satisfying $\begin{matrix}{{\frac{{fd} + {Bch}}{\left( {n + 1} \right){fsym}} \leqq k \leqq \frac{{fd} - {Bch}}{n\quad{fsym}}}{and}} & {{Exp}.\quad 12} \\{k \leqq \frac{fd}{2{fsym}}} & {{Exp}.\quad 14}\end{matrix}$ and N is an integer satisfying $\begin{matrix}{{\log\quad 2\left\{ \frac{{fd} + {Bch}}{\left( {n + 1} \right){Bch}} \right\}} \leqq N \leqq {\log\quad 2\left\{ \frac{2\left( {{fd} - {Bch}} \right)}{n\quad{Bch}} \right\}}} & {{Exp}.\quad 22}\end{matrix}$ where n is an integer satisfying $\begin{matrix}{1 \leqq n \leqq \frac{{fd} - {Bch}}{2{Bch}}} & {{Exp}.\quad 7}\end{matrix}$
 21. The wireless digital receiver according to claim 19,wherein the demodulation digital circuit includes: a quadraturedemodulator operable to quadrature-demodulate the signal oversampled bythe sampler; an automatic frequency controller operable to correct thesignal quadrature-demodulated by the quadrature demodulator to a signalhaving a component whose frequency is fi [Hz]; a low-pass filteroperable to low-pass-filter the signal frequency-corrected by theautomatic frequency controller; and a received data reproducing sectionoperable to reproduce received data from the signal low-pass filtered bythe low-pass filter.
 22. The wireless digital receiver according toclaim 19, wherein the frequency fj [Hz] is 3.000 [MHz].
 23. Anintegrated circuit for use in a wireless digital receiver in a wirelesscommunications system for transmitting/receiving a first wireless signalfrom a first wireless communications device and a second wireless signalfrom a second wireless communications device, the first and secondwireless signals having different frequency bands from each other, thewireless digital receiver receiving the second wireless signal in thefirst wireless communications device and digitally demodulating thesecond wireless signal, the integrated circuit comprising: a frequencyconversion section operable to downconvert the second wireless signaltransmitted from the second wireless communications device to alow-frequency signal; a sampling section operable to oversample thelow-frequency signal downconverted by the frequency conversion section;and a demodulation digital section operable to demodulate the signaloversampled by the sampling section, wherein: the signal demodulated bythe demodulation digital circuit has a center frequency of fi [Hz]; asampling frequency used in the sampling section and that used in thesecond wireless communications device are the same sampling frequency fs[Hz]; the sampling frequency fs [Hz] is set to a value that is aneven-number multiple of a wireless symbol transmission rate such thatoversampling is done in the sampling section and undersampling is donein a sampler of the second wireless communications device; and thecenter frequency fi [Hz] is ½ to 1 times a frequency corresponding to abandwidth of the first and second wireless signals and is ½^(N) (N is anatural number) times the sampling frequency fs [Hz].
 24. An integratedcircuit for use in a wireless digital receiver in a wirelesscommunications system for transmitting/receiving a first wireless signalfrom a first wireless communications device and a second wireless signalfrom a second wireless communications device, the first and secondwireless signals having different frequency bands from each other, thewireless digital receiver receiving the first wireless signal in thesecond wireless communications device and digitally demodulating thefirst wireless signal, the integrated circuit comprising: a frequencyconversion section operable to downconvert the first wireless signaltransmitted from the first wireless communications device to alow-frequency signal whose center frequency fd [Hz] is equal to adifference between a center frequency of the first wireless signal andthat of the second wireless signal; a sampling section operable toundersample the low-frequency signal downconverted by the frequencyconversion section; and a demodulation digital section operable todemodulate the signal undersampled by the sampling section, wherein: asampling frequency used in the sampling section and that used in thefirst wireless communications device are the same sampling frequency fs[Hz]; and the sampling frequency fs [Hz] is set to a value that is aneven-number multiple of a wireless symbol transmission rate such thatundersampling is done in the sampler and oversampling is done in asampler of the first wireless communications device.